CROSS REFERENCE TO RELATED APPLICATIONS

This is the U.S. National Stage of International Application No. PCT/GB2018/053133, filed Oct. 30, 2018, which was published in English under PCT Article 21(2), which in turn claims the benefit of Great Britain Application No. 1717857.5, filed Oct. 30, 2017.

The invention relates to amplifiers, in particular to low-noise amplifiers (LNAs) and more particularly to low-noise amplifiers employing reactive components as part of the impedance and/or noise-matching and gain-boosting mechanism.

There is a general desire to improve gain in amplifiers, especially in low-power RF receivers. However, as well as providing gain, it is also important that the LNA has a well-defined input impedance, so that it is power matched to the antenna for maximum power transfer between these two blocks in RF front-end. It will be appreciated that this is particularly important in LNAs working with already very low-levels input signals.

A basic common-gate (CG) LNA is shown inFIG. 2. The basic amplifying element of the amplifier200is a transistor M1which is biased in saturation by DC current source IDC. Active gain is simply provided by the transconductance, gmof the transistor M1, being defined as the ratio of output current to input voltage (iout/vin). As shown inFIG. 2, the gate of M1is AC grounded and kept at a constant DC voltage, while the input signal RFiis applied at the source, thus varying the gate-source voltage, and thereby producing a drain-source current. The output signal RF0is taken from the drain. For a CG-stage, output current is approximately equal to input current (i.e., intrinsic current gain is ≅=1). Thus, the input impedance for a CG-stage is defined as:

Zi=vi⁢nii⁢n=vi⁢nio⁢u⁢t≅1gm

To set the input impedance at a certain value for impedance matching, M1must be selected to have an intrinsic transconductance of a specific value, thus also determining the gain (product of gmand the load impedance, ZL) of the amplifier. This puts a limit on the gain of the amplifier. For example, to obtain an input impedance of 50Ω (typical impedance of an RF antenna), M1must have a transconductance of 20 milliamperes per volt (mA/V).

A modification of the CG-LNA ofFIG. 2uses a transformer to boost the transconductance, gmof the CG-stage, and with it the gain of the amplifier. This modification is shown inFIG. 3. The amplifier300has the same basic arrangement as inFIG. 2, except that an inverting transformer310is provided having a primary winding Lpand a secondary winding Ls. The primary winding Lpis connected to the source of transistor M1, and the secondary winding Lsis connected to the gate in inverting relationship such that the primary winding Lpsenses the input voltage, and the secondary winding Lsapplies an inverted and proportional voltage to the gate. Therefore, as the input voltage drops, the gate voltage increases, and vice versa. Thus, the gate-source voltage is passively amplified by the transformer300which in turn, combined with the intrinsic transconductance of M1, results in a larger overall transconductance. The overall gain depends on the characteristics of the transformer300, in particular its turns ratio, n and its coupling coefficient, k. For an ideal transformer, the coupling coefficient would be 1, but in practice will always be less than 1, typically 0.7-0.9 (moderate-to-strong mutual coupling). The input impedance of this amplifier300is defined as:

Zi≅1gm⁡(1+nk)

The transconductance of this amplifier300is improved by a factor (1+nk). However, from an impedance matching point of view, in order to match a specific impedance, such as a 50Ω antenna, it is still necessary to specifically set the intrinsic transconductance of the transistor M1at a specific level. Comparing the amplifier200ofFIG. 2with the amplifier300ofFIG. 3, the gm-boosted amplifier300can use a smaller transistor M1with lower transconductance, and/or can save power with a lower DC current. However, the circuit300is still constrained by the input impedance such that there is no overall improvement in signal gain. As an example, for an ideal transformer310(with n=1 and k=1), to match an input impedance of 50Ω, the intrinsic transconductance of M1, gm, has to be reduced from 20 mA/V to 10 mA/V. The increased passive gain from the transformer310has to be exactly offset by a reduced transconductance of transistor M1.

Another amplifier400is shown inFIG. 4. In this circuit the amplifier400uses a transformer410having a primary winding Lpand a secondary winding Lsin a positive (current) feedback arrangement. The primary winding Lpof the transformer410is connected to the drain of transistor M1and the secondary winding Lsis connected to the source. The primary winding Lpsenses the current at the output (drain) and applies this (non-inverted) at the input (source) in parallel to the input signal, thereby increasing the current through the transistor M1, and thus increasing the overall current gain of the amplifier400. The input impedance of this amplifier400is defined as:

Zi≅1gm⁡(1-kn)
where k is the coupling coefficient, n is the turns ratio, and k/n is the effective turns ratio of the transformer. While the overall current gain of the amplifier400is greatly improved (a much higher current gain can be obtained for a given input voltage), when viewed from an impedance matching perspective the circuit400again has limitations. This time, the factor (1−k/n) in the above equation means that for a given gm, the input impedance is increased. To compensate for this, and thereby achieve impedance matching, the intrinsic transconductance of the transistor M1has to be increased. As an example, for a transformer with n=1 and k=0.9, to match an input impedance of 50Ω, the intrinsic transconductance of M1, gmhas to be set at 200 mA/V. Thus, while the amplifier400results in a higher current gain (10), the increased transconductance requires higher power consumption for impedance matching, which is far from ideal, especially in applications for portable and/or battery powered devices.

According to the invention there is provided an amplifier, comprising:an amplifying element, having a voltage input across a first terminal and a third terminal and a voltage controlled current path between a second terminal and the third terminal; anda trifilar transformer having a primary winding, a secondary winding and a tertiary winding;wherein the primary winding is connected to the third terminal, the secondary winding is connected to the first terminal and the tertiary winding is connected to the second terminal;wherein the primary winding and the secondary winding are mutually coupled in inverting relationship;wherein the primary winding and the tertiary winding are mutually coupled in non-inverting relationship;wherein the secondary winding and the tertiary winding are mutually coupled in inverting relationship; andwherein the tertiary winding is between the amplifier output and the second terminal.

The trifilar transformer arrangement provides more than one gain mechanism by which the overall transconductance/gain of the circuit is increased. Firstly, the voltage feedforward arrangement provided by the primary winding and the secondary winding increases the gain by sensing the voltage at the amplifier input on the third terminal and applying it with inversion to the voltage input at the first terminal. This increases the voltage across the first and third terminals, which in turn controls the magnitude of the current through the voltage controlled current path, thereby increasing the transconductance of the amplifying element. At the same time, the positive current feedback provided by the mutual coupling of the tertiary winding and the primary winding causes a second gain mechanism which increases current flow through the voltage controlled current path, and thus also provides increased current gain of the amplifier.

A third gain mechanism is also in operation due to the mutual coupling of the tertiary winding and the secondary winding. The voltage sensed by the tertiary winding is coupled in positive feedback arrangement to the voltage input on the first terminal. However, it should be noted that this third gain mechanism is only desirable if it can be kept at a low enough level. If the mutual coupling between the secondary and tertiary windings is too high, then there is a risk of instability and the circuit oscillating. However, at a sufficiently low level, this third gain mechanism is beneficial, providing added overall gain. More specifically, in order to ensure stability of the amplifier when its gain is greater than or equal to 1 (in the frequency range from dc up to the transit frequency), the effective turns ratios of each pair of windings (i.e. n1/k1, n2/k2and n3/k3) should be chosen such that the real part of the input impedance is positive and the real part of the output impedance is positive. This may be achieved by selecting the effective turns ratios of each pair of windings such that the phase difference between the first terminal and the third terminal (gate and source in the case of a single transistor) is within a range of 120-240 degrees (i.e. 180+/−60 degrees), preferably within a range of 150-210 degrees (i.e. 180+/−30 degrees).

In some embodiments the trifilar transformer may be arranged such that the mutual coupling between the secondary winding and the tertiary winding is lower than the mutual coupling between the primary winding and the secondary winding and/or lower than the mutual coupling between the primary winding and the tertiary winding. By keeping the secondary-tertiary mutual coupling lower than that of the other windings, the amplifier is more likely to be stable (although stability is not guaranteed across all choices of ni, ki). The coupling coefficients of the various windings can be adjusted by transformer design, e.g. by adjusting the relative sizes and/or positioning of the windings.

Thus, the amplifier provides an excellent overall gain. However, an additional and significant benefit of this arrangement is that the input impedance of the circuit can be adjusted for impedance matching without adversely impacting the gain or putting undesirable constraints on the intrinsic transconductance of the amplifying element. The input impedance of the amplifier is defined as:
Zi≅(gm(1+nP,SkP,S+nT,SkT,S)(1−kT,P/nT,P))

where:nP,Sis the turns ratio of the primary and secondary windings;nT,Sis the turns ratio of the tertiary and secondary windings;nT,Pis the turns ratio of the tertiary and primary windings;kP,Sis the mutual coupling coefficient of the primary and secondary windings;kT,Sis the mutual coupling coefficient of the tertiary and secondary windings;kT,Pis the mutual coupling coefficient of the tertiary and primary windings;gmis the intrinsic transconductance of the amplifying element.

In the above equation, it can be seen that there is a factor introduced by the transformer which is dependent on the three mutual couplings. Advantageously, this factor is the product of two terms. The first term, (1+nP,SkP,S+nT,SkT,S) is always greater than 1 and the second term, (1−kT,P/nT,P) is always less than 1 (although care should be taken that this second term remains positive, which will for example always be the case where n is greater than or equal to 1). Therefore, by careful selection of the turns ratios and coupling coefficients the input impedance can be matched to a particular value, while still having excellent gain and without having to use an amplifying element with particularly large transconductance, and thus, power consumption.

In some preferred embodiments the primary winding is mutually coupled to each of the secondary winding and the tertiary winding, and the secondary winding and the tertiary winding are substantially not coupled to each other. This arrangement is preferred where the secondary-tertiary coupling is entirely unwanted and should be minimized or entirely eliminated. However, as discussed above, in many applications some secondary-tertiary coupling can be tolerated and is even beneficial. As discussed above, although not a guarantee of amplifier stability, a reduced mutual coupling coefficient between the secondary and tertiary windings is a good indicator in many embodiments. Therefore, in some preferred embodiments the mutual coupling coefficient between the secondary and tertiary windings is less than the mutual coupling coefficient between the primary and secondary windings and/or less than the mutual coupling coefficient between the primary and tertiary windings. More preferably the mutual coupling coefficient between the secondary and tertiary windings is less than two thirds, preferably less than half, more preferably less than one third of the mutual coupling coefficient between the primary and secondary windings and/or the mutual coupling coefficient between the primary and tertiary windings.

Perfect coupling between the primary winding and each of the secondary and tertiary winding is normally not achievable, especially in on-chip transformers where the size and shape of the windings are constrained by the fabrication processes, but strong coupling can be obtained, e.g. a coupling coefficient of around 0.8-0.9 may be possible in some embodiments. The magnitude of the mutual coupling coefficient between the secondary winding and the tertiary winding that may be used with a stable amplifier will depend on the particular circuit arrangement, but in some particularly preferred embodiments the mutual coupling coefficient between the secondary winding and the tertiary winding is less than 0.4, preferably less than 0.3, more preferably less than 0.2, yet more preferably less than 0.1.

The three windings of the trifilar transformer may be arranged in any of a number of different configurations. For example, the windings may be concentric, interwound or stacked or any combination of these. For on-chip transformers, the windings are formed in thick (or ultra-thick) metal layers, and in such arrangements the windings may all be in a single layer, two stacked layers or even three stacked layers. In some particularly preferred embodiments the primary winding, the secondary winding and the tertiary winding are all concentric and the primary winding separates the secondary winding from the tertiary winding. By placing the primary winding in between, the secondary winding and the tertiary winding, there will be stronger coupling between the primary and secondary and between the primary and tertiary than there will be between the secondary and tertiary as the secondary and tertiary windings have the greatest separation.

In other embodiments the primary winding may be interwound with either the secondary winding or the tertiary winding. The other of the secondary winding and the tertiary winding may be concentric with the interwound windings.

In other embodiments the trifilar transformer may be formed in two metal layers with the primary winding in a different layer from one of the secondary winding and the tertiary winding and the primary winding being formed in the same layer and concentric with the other of the secondary winding and the tertiary winding.

As discussed above, in some cases it may be desirable to reduce the mutual coupling of the secondary and tertiary windings to the maximum extent possible, practically to zero if possible. This may be achieved by appropriate shaping of the windings. Therefore, in some preferred embodiments the secondary winding is shaped so as to have near-zero mutual coupling with the tertiary winding. This will of course depend on the relative shapes of both the secondary and tertiary windings. Near-zero mutual coupling may be considered in relative terms when compared with the mutual coupling of the other winding pairs (e.g. less than a fifth of those). However, by way of example in some embodiments, e.g. where the other winding pairs have mutual couplings of around 0.5-0.7, near-zero mutual coupling may be taken to mean a mutual coupling of less than 0.1, preferably less than 0.05. One configuration that cancels the coupling is where one winding is a loop winding and the other is a figure-of-eight winding such that the coupling along one half of the ‘eight’ cancels the coupling along the other half of the ‘eight’. A third winding can be arranged to couple with both of these windings. This arrangement is beneficial in differential amplifier embodiments (when driven symmetrically, impedances are the same on the two ports of each winding).

While it will be appreciated that the following is not limiting on the invention, in some preferred embodiments the amplifying element has an intrinsic transconductance of between 10 mA/V and 100 mA/V, preferably between 20 mA/V and 50 mA/V. With the architecture described here, the amplifier can obtain high gain while keeping the intrinsic transconductance of the amplifying element within these more normal and optimal ranges while also achieving impedance matching.

Although it will be appreciated that the principles described above may apply to any amplifying means, the amplifying element is preferably a transistor. For example, the amplifying element may be a number of transistors or other components connected together to form an amplifying circuit. However, in certain preferred embodiments a single transistor is used as the amplifying element. The circuit described here is particularly advantageous in providing a simple amplifier with high gain and impedance matching in a simple circuit with a single transistor as the main amplifying element. The transistor may be any type of transistor such as a bipolar junction transistor (BJT), but more conveniently the transistor may be a FET, preferably a MOSFET.

The FET is preferably arranged in a common-gate configuration. It will be appreciated that in a BJT arrangement, this equates to a common-base configuration.

In particularly preferred embodiments where the amplifying element is a FET, the primary winding is connected to the FET's source, the secondary winding is connected to the FET's gate and the tertiary winding is connected to the FET's drain.

According to another aspect, the invention provides a method of amplifying a signal with an amplifying element, the amplifying element comprising a voltage input across a first terminal and a third terminal and comprising a voltage controlled current path between a second terminal and the third terminal, the method comprising:applying the signal to a third terminal of the amplifying element;sensing, with a primary winding of a trifilar transformer, a voltage at the third terminal;coupling with inversion at least a part of said sensed voltage from the third terminal via a secondary winding of the trifilar transformer to the first terminal of the amplifying element;sensing, with a tertiary winding of a trifilar transformer, a current at the second terminal;coupling without inversion at least a part of said sensed current from the second terminal via a tertiary winding of the trifilar transformer to the third terminal of the amplifying element; andoutputting an amplified signal from an output node located such that the tertiary winding is between the output node and second terminal.

It will be appreciated that all of the preferred and optional features discussed above may also be applied correspondingly to the method of operation.

