to understand the motivation of the proposed power allocation method and the substantial performance improvement that can be achieved with it over wireless channels , some numerical examples are now described . the examples are shown in fig1 to 4 . these figures represent the abep of a n t × 1 sm - mimo system , with n t denoting the number of transmitter elements at a transmitter . throughout this application , when the term “ transmitters ” or “ transmit - antenna ” is used , this refers to transmitter elements at the transmitter . accordingly , the term “ receivers ” or “ receive - antenna ” refers to receiver elements . fig1 to 4 are obtained by considering a nakagami - m fading channel with parameters { m i } i = 1 n t and { ω i } i = 1 n t on a wireless link between an i - th transmit - antenna and a single receive - antenna . fig1 to 4 are obtained by assuming a uniform power allocation mechanism among the transmit - antennas ( e m / n 0 denotes the average signal - to - noise ratio , snr , for each wireless link ), but for balanced ( i . e ., { ω i } i = 1 n t are the same in all wireless links ) and unbalanced ( i . e ., { ω i } i = 1 n t can be different among the various wireless links ) fading channels . it will be described in further detail below that the latter system setup is equivalent to have balanced fading channels with a non - uniform power allocation among the transmit - antennas . fig1 to 4 show both monte carlo simulations and an accurate analytical framework . the analytical model will be described in more detail below / with regard to fig5 . in fig1 , the abep is shown when a fading correlation model is considered . the following scenarios are applied : scenario a : f 1 = f 2 = 1 , ω 1 = ω 2 = ⅔ , ρ 1 = 0 . 50 , ρ 2 = ρ 3 = ρ 4 = 0 . 45 . scenario b : f 1 = 1 , f 2 = 2 . 5 , ω 1 = 1 , ω 2 = 20 / 3 , ρ 1 = 0 . 50 , ρ 2 = ρ 3 = ρ 4 = 0 . 45 . scenario c : f 1 = f 2 = 1 , ω 1 = ⅔ , ω 2 = 20 / 3 , ρ 1 = 0 . 50 , ρ 2 = ρ 3 = ρ 4 = 0 . 45 . n t = 2 . both monte carlo simulation ( markers ) and analytical model ( solid lines ) are shown . the abep changes significantly with the correlation coefficient ( ρ 1 , ρ 2 , ρ 3 , ρ 4 ) and the fading severity ( f 1 , f 2 ). this latter phenomenon can be observed by comparing scenario b and scenario c in fig1 , where a non - negligible performance gap is observable ( for the same power imbalance ratio ω 2 / ω 1 ). this result emphasizes that the performance of sm is strongly affected by the characteristics of the wireless channel and the fading distribution as well , since scenario c is representative of a rayleigh fading channel . in fig2 , the abep is shown for uncorrelated fading channels and for a different power imbalance among the wireless links . for the balanced fading channel , ω 1 = ω 1 = ω 1 = ω 1 = 1 is applied . for the unbalanced fading channel , ω 1 = 1 , ω 1 = 4 , ω 1 = 8 , ω 1 = 12 is applied . the fading channels are uncorrelated , f 1 = f 2 = f 3 = f 4 = 2 . 5 and n t = 4 . both monte carlo simulation and analytical model are shown . a significant performance improvement can be observed when the power links are unbalanced . the reason for this behaviour lies in the fact that power imbalance makes the wireless links more distinguishable to each other , thus making the decision process at the receiver more reliable and , thus , improving the abep . in fig3 , the abep is shown for correlated fading channels and for a different power imbalance among the wireless links . the following scenarios are applied : further , ρ 1 = ρ 2 = ρ 3 = 0 . 45 , σ 4 =− 0 . 45 is applied and n t = 2 . both monte carlo simulation and analytical model are shown . similar to fig2 , a better performance can be achieved for unbalanced fading channels . furthermore , an interesting comment can be made by carefully observing , in particular , the curves related to scenario b and scenario c . in both system setups the first wireless link has a greater power gain , but in scenario b the fading severity ( f 1 ) of the first link is smaller than the fading severity ( f 2 ) of the second link . a significant performance difference , which highlights that besides the average power also the fading severity can remarkably alter the system performance . in fig4 , the abep for correlated fading is shown . for the balanced fading channel , ω 1 = ω 2 = ω 3 = ω 4 = 1 is applied . for the unbalanced fading channel , ω 1 = 1 , ω 2 = 4 , ω 3 = 8 , ω 4 = 12 is applied . the fading channels are correlated with the correlation coefficient ρ i , j = exp (− d 0 | i − j |) with d 0 = 0 . 22 and ( i , j ) are antenna &# 39 ; s indexes ( i , j = 1 , 2 , 3 , 4 ). n t = 4 . both monte carlo simulation and analytical model are shown . the results in fig4 confirm the conclusions already drawn in fig1 to 3 , but also highlight that the fading severity ( f ) can have a different impact on the abep , as far as balanced and unbalanced system setups are considered : a different relation among the curves is noticed . in summary , the numerical results shown in fig1 to 4 lead to the following conclusions : 1 . the abep of sm gets better for unbalanced wireless fading channels . 2 . the abep of sm depends on the fading severity and channel correlation of the wireless links . according to an aspect of the present invention , abep - driven opportunistic power allocation mechanisms are described that aim at distributing the available power at the transmitter in an optimal fashion by taking into account the actual characteristics of all transmit - receive wireless links . in particular , the method according to this aspect of the present invention will help to make the wireless links more distinguishable among each other in order to emulate unbalanced fading conditions for those scenarios where the channel fading is actually identically distributed . in fig5 , a system 10 for sm - mimo is shown by way of example , which is composed by two transmit - antennas 12 , 14 ( tx 1 , tx 2 ) at the transmitter 16 ( tx ) and one receive - antenna 18 at the receiver 20 ( rx ). the system 10 is used to describe the basic and fundamental idea behind the proposed power allocation method herein below . for analytical simplicity , a rayleigh fading channel model with correlated channels 22 , 24 is considered . however , it is emphasized that the invention outlined herein is applicable to generic mimo and miso systems that can be deployed in any wireless fading channel . the main aim is to show that power imbalance and non - uniform power allocation are equivalent to each other . a complex - envelope representation of signals is used throughout the description . j =√{ square root over (− 1 )} is the unitary unit . ( x y )( t )=∫ − x + x x ( ξ ) y ( t − ξ ) dξ is the convolution operator of signals x (•) and y (•). (•)* denotes complex - conjugate . ∥ 2 denotes the square absolute value . e {•} denotes expectation operator . re {•} denotes real part operator . pr {•} means probability . g ˜ n ( μ g , μ g 2 ) is a gaussian distributed random variable ( rv ) with mean μ g and standard deviation σ g . a ˜ r (•; σ g 2 ) is a raleigh distributed rv with e { a 2 }= 2σ a 2 p ab denotes the correlation coefficient of rvs a and b . q ( x )=( 1 /√{ square root over ( 2π )} m )∫ x + x exp (− t 2 / 2 ) dt is the q - function . γ ( x )=∫ 0 + x t x − 1 exp (− t ) dt is the gamma function . m 1 and m 2 denote the two information messages that the transmitter ( tx ) in fig5 can emit . { circumflex over ( m )} denotes the message estimated at the receiver . e m 1 = ζ 1 e m and e m 2 = ζ 2 e m are the energies transmitted for the information messages m 1 and m 2 , respectively . ζ 1 and ζ 2 are power scaling variables . e m is a constant factor . t m = t m 1 = t m 2 denote the signalling internal for both information messages m 1 and m 2 . h 1 ( t )= β 1 exp ( jφ 1 ) δ ( t − τ 1 ) is the channel impulse response from antenna tx 1 to the receive - antenna , and β 1 , φ 1 , and τ 1 denote the gain , phase , and delay of the related wireless link . moreover , α 1 = β 1 exp ( jω 1 ) denotes the channel &# 39 ; s complex gain of the first wireless link . analogously , h 2 ( t )= β 2 exp ( jφ 2 ) δ ( t − τ 2 ) is the channel impulse response from antenna tx 2 to the receive antenna , and α 2 , φ 2 , and τ 2 denote the gain , phase , and delay of the related wireless link . moreover , α 2 = β 2 exp ( jφ 2 ) denotes the channel &# 39 ; s complex - gain of the second wireless link . for the sake of simplicity , but without loss of generality , a rayleigh fading is assumed for both wireless links tx 1 rx and tx 2 - rx . in particular , α 1 = α 1 r + jα 1 1 and α 2 = α 2 r + jα 2 1 with α 1 r ˜ n ( 0 , σ 1 2 ), α 1 1 ˜ n ( 0 , σ 1 2 ), α 2 r ˜ n ( 0 , σ 2 2 ), α 2 1 ˜ n ( 0 , σ 2 2 ) and ρ α 1 r α 1 1 = ρ α 2 r α 2 1 = ρ α 1 r α 2 1 = ρ α 2 3 α 1 1 = 0 , ρ α 1 r α 2 r = ρ α 1 α 2 1 = ρ . τ 1 and τ 2 are assumed to be independent and uniformly distributed in [ 0 , t m ], but known at the receiver , i . e . perfect time - synchronization is considered . the receiver is assumed to have full channel state information ( csi ), i . e . two triples ( β 1 , φ 1 , τ 1 ) and ( β 2 , φ 2 , τ 2 ) are perfectly known at the receiver . csi can be obtained during a training phase before data transmission . the signals transmitted by antennas tx 1 and tx 2 are denoted by s 1 (•) and s 2 (•), respectively , which after passing through the wireless channel becomes { tilde over ( s )} 1 ( t )=( s 1 { circle around ( x )} h 1 )( t )= β 1 exp ( jφ 1 ) s 1 ( t − τ 1 ) and { tilde over ( s )} 2 ( t )=( s 2 { circle around ( x )} h 2 )( t )= β 2 exp ( jφ 2 ) s 2 ( t − τ 2 ), respectively . the noise at the receiver input is denoted by n (•), and is assumed to be additive white gaussian ( awg ) distributed , with both real and imaginary parts having a double - sided power spectral density equal to n 0 . the receiver signal is denoted by r (•), and is equal to r ( t )={ tilde over ( s )} 1 ( t )+{ tilde over ( s )} 2 ( t )+ n ( t ). for ease of notation , γ = e m /( 4n 0 ) is set . the spatial modulation ( sm ) concept is based on the rule as follows : i ) when message m 1 has to be transmitted , a properly designed signal s 1 ( t ) ≠ 0 is sent by only the transmit - antenna tx 1 ( i . e ., s 2 ( t )= 0 ), and ii ) when message m 2 has to be transmitted , a properly designed signal s 2 ( t )≠ 0 is sent by only the transmit - antenna tx 2 ( i . e ., s 1 ( t )= 0 ). in other words , only one transmit - antenna is activated when either m 1 or m 2 have to be sent : there is only one active transmit - antenna for each signalling time - interval t m . furthermore , the active transmit - antenna is also allowed to transmit modulated data and , as a consequence , information is conveyed by both a transmit - antenna index and the modulated signal transmitted by that transmit - antenna . this is incorporated in the power allocation method described herein . it is assumed that the transmitted signals , when different from zero , are pure sinusoidal tones , i . e ., s 1 ( t )=√{ square root over ( e m 1 )} exp ( jω c t ) and s 2 ( t )=√{ square root over ( e m 2 )} exp ( jω c t ). in such a case , we have : where , with a slight abuse of notation , both delays τ 1 and τ 2 have been embedded into the channel phases φ 1 and φ 2 , respectively . as a consequence , the received signal is : the maximum likelihood ( ml ) optimal detector with perfect channel knowledge and time - synchronization at the receiver is as follows : so , the probability of error p e (•,•) conditioned upon the channel impulse responses h 1 (•) and h 2 (•) is as follows : then , the abep over correlated rayleigh fading channels can be obtained as follows ( abep = e h 1 , h 2 { p e ( h 1 , h 2 )}): where we have defined σ 2 ={ tilde over ( σ )} 1 2 +{ tilde over ( σ )} 2 2 − 2ρ { tilde over ( σ )} 1 { tilde over ( σ )} 2 with { tilde over ( σ )} 1 2 = ζ 1 σ 1 2 and { tilde over ( σ )} 2 2 = ζ 2 σ 2 2 . the result shown in ( 8 ) takes into account an arbitrary power allocation between the two transmit - antennas , i . e ., in general , ζ 1 ≠ ζ 2 . a uniform power allocation strategy is obtained by setting ζ 1 = ζ 2 = 1 . the power imbalance between the transmit - receive wireless links is equivalent to non - uniform power allocation . this can be readily proven by considering the following facts : 1 . if α 1 and α 2 are rayleigh distributed rvs with probability density function ( pdf ) f α 1 ( ξ 1 )˜ r ( ξ 1 ; σ 1 ) and f α 2 ( ξ 2 )˜ r ( ξ 2 ; σ 2 ) as follows , respectively : 2 . then , { tilde over ( α )} 1 =√{ square root over ( ζ 1 )} α 1 and { tilde over ( α )} 2 =√{ square root over ( ζ 2 )} α 2 are still rayleigh distributed rvs with pdfs equal to f { tilde over ( α )} 1 ( ξ 1 )˜ r ( ξ 1 ;√{ square root over ( ζ 1 )} σ 1 ) and f { tilde over ( α )} 2 ( ξ 2 )˜ r ( ξ 2 ;√{ square root over ( ζ 2 )} σ 2 ), respectively . in the light of 1 . and 2 ., it follows that even though σ 1 = σ 2 , which corresponds to a balanced fading scenario , the net effect of a non - uniform power allocation , ζ 1 ≠ ζ 2 , in ( 8 ) is equivalent to have an unbalanced fading scenario with power imbalance ratio equal to e {{ tilde over ( α )} 1 2 }/ e {{ tilde over ( α )} 2 2 }= ζ 1 / ζ 2 . as a consequence , the abep depicted in fig1 to 4 for the balanced fading scenario can be moved towards the abep of the unbalanced fading scenario via an adequate and opportunistic ( i . e ., which depends also on the fading parameters — see , e . g ., fig3 ) power allocation scheme . to formalize the power allocation mechanism , a general system setup with n t transmit - antennas and one receive - antenna is considered , as shown for example in fig5 . further generalizations to an arbitrary number of receive - antennas are possible provided that closed - form and accurate expressions of the abep are computed . a general fading scenario with nakagami - m fading ( the same channel model considered in fig1 to 4 ) is also considered . for illustrative purposes , the power allocation optimization is formulated by considering the abep for uncorrelated fading channels and for a receiver having full channel state information ( csi ). however , the same method can be used for correlated fading channels and for other sub - optimal receiver architectures . in fig5 , 6 and 7 , a n , and a 1 and a 2 respectively , denote the power scaling variables . furthermore , c 1 , 2 denotes the correlation coefficient between the channels . the transmitter 16 comprises a spatial modulator 26 and a signal modulator 28 . at the receiver 20 , a signal demodulator 30 is located . the receiver also comprises a channel estimator 32 which determines correlation coefficients between the channels 22 , 24 . these correlation coefficients are transmitted to the transmitter 16 . the transmitter 16 comprises a calculator 34 to determine a power allocation to be allocated to the transmit - antennas 12 , 14 on the basis of data 35 comprising correlation coefficients measured at the receiver 20 . the transmitter 16 further comprises a power allocator 36 to allocate the determined power to the transmit - antennas 12 , 14 . a modulated signal is then transmitted from one transmit - antenna 12 , 14 at a time with the corresponding allocated power . at the receiver 20 , a model of power allocation is generated in a model generator 38 , in which the data 35 of the channel estimator 32 is applied . the location of the transmit - antennas 12 , 14 is then determined in a detector 40 located at the receiver 20 by using the power allocation model . in the depicted scenario , the abep can be written in closed - form as follows : and emphasized that the above abep is conditioned upon a fixed power allocation among the transmit - antennas , i . e ., the set ( ζ 1 , ζ 2 , . . . , ζ n t ). moreover , we have defined : being the meijer - g function , and a i = 2m i m i /({ tilde over ( ω )} i m i γ ( m i )), b i = m i /{ tilde over ( ω )} i . c i = 2m i − 1 , { tilde over ( ω )} 1 = ζ i ω i , for i = 1 , 2 , . . . , n t . moreover , ( m i , ω i ) are the parameters of the nakagami - m distribution for the i - th wireless link . the optimal and opportunistic power allocation mechanism based on the optimization of the abep can be analytically formalized as follows : where the constraint that the total power is fixed to a given value , i . e ., σ i = 1 n t e m i = e m has been added . the minimization in ( 14 ) turns out to be a constrained optimization problem , which can be solved via either analytical or numerical methods . in fig6 , a second embodiment of a system 100 according to the present invention is shown . a transmitter 102 is provided with two transmit - antennas 104 , 106 and further comprises a spatial modulator 108 and a signal modulator 110 . at a receiver 112 , a signal demodulator 114 is located . the receiver 112 has a receive - antenna 116 and also comprises a storage 118 for storing power allocation sequences , for example in a look - up table . the transmitter 102 comprises a sequence predetermination module 120 in which a power allocation sequence 122 , 124 to be allocated to the transmit - antennas 104 , 106 is predetermined . this power allocation sequence 122 , 124 is randomly chosen prior to any data transmission from the transmit - antennas to the receiver 112 and does not consider channel conditions . the transmitter 102 further comprises a power allocator 126 to allocate the predetermined power allocation sequences to the transmit - antennas 104 , 106 . a modulated signal is then transmitted from the transmit - antennas 104 , 106 , i . e . from the transmit - antenna 104 , 106 which is active at a certain time interval , with the corresponding power allocation sequence . the power allocation sequences 122 , 124 of two transmit - antennas , which are shown in more detail , are only an example of possible power allocation sequences . the power allocation sequences 122 , 124 are shown as a function of time ( t ). they have a staircase - shaped characteristic which differs for each of the transmit - antennas . for example , in a first time interval 128 , the power ( a 1 ) of the first transmit - antenna 104 is lower than the power ( a 2 ) of the second transmit - antenna 106 . then , in a second time interval 130 , the power ( a 1 ) of the first transmit - antenna 104 is higher than the power ( a 2 ) of the second transmit - antenna 106 . the power allocation sequences 122 , 124 of each transmit - antenna 104 , 106 are transmitted to the receiver 110 and stored in the storage 118 . the receiver 112 further comprises a detector 132 to detect location of the active transmit - antenna by using the power allocation sequence data stored in the storage 118 . fig7 shows a further embodiment of a system 101 according to the present invention by way of example . in this embodiment is similar to the embodiment shown in fig6 and therefore , the same reference signs are used for similar items . however , channel conditions are considered prior to the allocation of a power allocation sequence 122 , 124 to the transmit - antennas 104 , 106 . therefore , the receiver 112 further comprises a channel estimator 134 . training data 136 is transmitted from the receiver 112 to the transmitter 102 , containing channel information that has been analyzed in the channel estimator 134 . the channel information mainly comprises gain , phase and / or delay of the signals received at the receiver 112 from different transmit - antennas . the training data provides a feedback to the transmitter 102 such that the power allocation sequence 124 , 126 is randomly chosen based the channel information . the predetermined power allocation sequence 124 , 126 is transmitted to the receiver 112 again to be stored in the storage 118 which is accessible for the detector 132 to determine the location of the active transmit - antenna . with regard to the embodiments shown in fig6 and 7 , it is also possible that a constant power ( a 1 , a 2 ) is allocated to each of the transmit - antennas 104 , 106 , such that a 1 ≠ a 2 . the difference between these constant powers can depend on the channel correlation , for example , the difference can be direct proportional to the channel correlation . another aspect of the invention focuses on proposing a novel spatial modulation method , which is called time - orthogonal signal design assisted spatial modulation ( tosd - sm ) and , differently from all other sm techniques , can exploit the antennas at the transmitter to get transmit - diversity . the basic idea behind tosd - sm is not restricting the transmitted signal to be a pure sinusoidal tone , but to properly design it in order to exploit , in an efficient way , the different propagation delays ( τ 1 and τ 2 ) of the wireless links tx 1 - rx and tx 2 - rx . similar to the first aspect of the invention , tosd - sm retains the main assumption that only one transmit - antenna is activated for every signalling interval t m . in particular , also in this aspect , the following rule is adopted : i ) when message m 1 has to be transmitted , a properly designed signal s 1 ( t )≠ 0 is sent by only the antenna tx 1 ( i . e ., s 2 ( t )= 0 ), and ii ) when message m 2 has to be transmitted , a properly designed signal s 2 ( t )≠ 0 is sent by only the antenna tx 2 ( i . e ., s 1 ( t )= 0 ). the assumption that only one transmit - antenna is activated for every signalling interval t m can also be used for more than two antennas . tosd - sm does not restrict the transmitted signals to be pure sinusoidal tones , but s 1 (•) and s 2 (•) are properly optimized for performance improvement . in particular , the novel tosd - sm concept relies on the following signal design ( when s 1 (•) and s 2 (•) are different from zero ): where w (•) is a generic signal waveform , which is chosen to satisfy the following co the design condition in ( 19 ) simply states that w (•) is required to have a very peaky time auto - correlation function r w (•), which under ideal signal design conditions can be assumed to be a dirac &# 39 ; s delta function . according to ( 18 ), the signals after propagation through the wireless channels can be written as follows : the ml optimal detector with perfect channel knowledge and synchronization at the receiver is as follows : similar to ( 5 ), the probability of error p e (•,•) conditioned upon the channel impulse responses h 1 (•) and h 2 (•) is as follows : after some analytical calculations and the exploitation of the orthogonality condition in ( 19 ) for every pair of delays ( τ 1 , τ 2 ) with τ 1 ≠ τ 2 , ( i . e ., propagation through the wireless links tx 1 - rx and tx 2 - rx is subject to different delays ), i . e . : then , p e . over rayleigh fading channels can be obtained as follows : where we have defined m ( s )=[ 1 + 2 ( σ 1 2 + σ 2 2 ) s + 4 ( 1 − ρ 2 ) σ 1 2 σ 2 2 s 2 ] − 1 , which is the moment generating function ( mgf ) of β = β 1 + β 2 , i . e ., m ( s )= e { exp (− sβ )}. the main advantage of this aspect of the invention is to provide transmit - diversity . in particular , for a 2 × 1 miso system a transmit - diversity order equal to 2 is obtained . in particular , the diversity order can be computed by analyzing the behaviour of m (•) for large values of | s |. it can be readily proven that : and it is known that the system &# 39 ; s diversity order is equal to the negative exponent of | s |, i . e ., 2 in ( 29 ). as a result of the higher diversity order , the error probability is expected to have a steeper slope for increasing snrs , which results in substantial improvements in system &# 39 ; s performance . this aspect of the invention turns out to be also more robust to channel correlation . as a matter of fact , the error probability in ( 28 ) depends on only the square value of the correlation coefficient , i . e ., ρ 2 . so , since 0 ≦ ρ ≦ 1 the performance drop for increasing ρ is expected to be smaller than in known solutions . as opposed to known solutions in which the performance of the sm scheme is independent of ρ , a transmit - diversity is achieved . with respect to other sm schemes , it might be required that the propagation delays ( τ 1 , τ 2 ) are known at the transmitter . when needed , this can be easily obtained via a feedback channel from the receiver to the transmitter , such that the orthogonality condition in ( 25 ) can always be verified . on the other hand , when the signal design condition in ( 19 ) can be guaranteed , a priori , for every ( τ 1 , τ 2 ) pair , no feedback channel is required since the condition in ( 25 ) is implicitly verified for every pair ( τ 1 , τ 2 ) at the receiver - side . in addition to or instead of propagation delays , other channel signatures ( or spectral characteristics ) such as phase rotation , amplitude or frequency may be used . below , numerical results which are obtained from the analytical frameworks described above , are described . the following system setup is used to obtain the simulation results : iv ) the error probability from monte carlo simulations is obtained by requiring a number of wrong detections equal to 10 4 . numerical results are shown in fig8 , fig9 , and fig1 for known sm schemes and the proposed tosd - sm , respectively . markers show a monte carlo simulation and solid lines show the analytical model . in particular , fig8 shows the error probability of a sm scheme which is called space shift keying ( ssk ) and based on the rule that only one transmit - antenna is activated when m 1 has to be sent , while both transmit - antennas are activated when m 2 needs to be sent . in this scheme , the transmitted signals , when different from zero , are always pure sinusoidal tones . this allows to embed both delays ( τ 1 and τ 2 ) into the channel phases φ 1 and φ 2 , respectively . numerical results confirm that no performance degradation can be observed for increasing values of the correlation coefficient . in fig9 , the error probability of another sm scheme , based on the rule that only one transmit - antenna is activated when either m 1 or m 2 have to be sent : there is only one active transmit - antenna for every signaling interval t m . similar to the ssk scheme the transmitted signals , when different from zero , are always pure sinusoidal tones , i . e ., s 1 ( t )= s 2 ( t )=√{ square root over ( e m )} exp ( jω c t ). thus , also in this case both propagation delays τ 1 and τ 2 can be embedded into the channel phases φ 1 and φ 2 , respectively . it is observed that the spatial correlation between the wireless links can remarkably increase the error probability . when the wireless links are subject to high correlation , the error probability can be very high . more in detail , the snr penalty with respect to spatial correlation is 1 . 24 db , 3 db , and 6 db for ρ = 0 . 25 , ρ = 0 . 5 , and ρ = 0 . 75 , respectively . in fig1 , the error probability for the novel tosd - sm scheme is depicted . numerical results confirm that the proposed invention shows a higher diversity order than other sm schemes : the error probability shows a steeper slope than the other two proposals . this yields a substantial performance gain with respect to other solutions . moreover , we can observe that spatial correlation of wireless links has a significant less impact than the sm proposal as described in relation with fig9 . in fig1 a comparison among the various sm proposals is shown in order to understand the different behaviour of them as a function of channel spatial correlation . the following facts can be observed : i ) the proposed tosd - sm yields a significant performance gain with respect to all other sm proposals and , even in the presence of channel correlation , it offers better error probabilities than other sm schemes over independent wireless links : this is a clear indication of the robustness of the proposed invention to spatial correlation of fading . ii ) the sm proposal referring to space shift keying offers worse performance than the sm scheme of which results are shown in fig9 when the wireless links are uncorrelated . however , in the presence of channel correlation the situation is reversed : sm for ssk offers a better error probability than for the sm scheme which results are shown in relation to fig9 . fig1 shows a system 200 for sm - mimo , in accordance with an embodiment of the tosd - sm scheme . the system 200 comprises a transmitter 202 with two transmit - antennas 204 , 206 and a receiver 208 with one receive - antenna 210 . the transmitter 202 comprises a spatial modulator 212 and a signal modulator 214 . the receiver 208 comprises a signal demodulator 216 for demodulating a signal which is sent over a channel 218 , 220 from one transmit - antenna 204 , 206 to the receive - antenna 210 . each signal transmitted over the channels 218 , 220 has a specific waveform . according to the tosd - sm scheme , the waveform of the signal is predetermined prior to any data transmission between the transmitter 202 and the receiver 208 such that a propagation delay of the signals is modified . corresponding data 222 , which comprise orthogonal pairs ( τ 1 , τ 2 ) of the propagation delay between the two channels 218 , 220 are supplied to a waveform calculator 224 located at the transmitter 202 . the predetermined waveform is then allocated to the corresponding signal sent by the transmit - antennas 204 , 206 via a waveform allocator 226 located at the transmitter 202 . the receiver 208 comprises a database 228 in which a model of waveforms is provided . the receiver 208 further comprises a detector 230 which uses knowledge of the conditions of the channels 218 , 220 and the waveform model to detect location of the active transmit - antenna 204 , 206 . furthermore , a time - synchronisation at the receiver 208 is considered . the resulting time - orthogonal design of the signals reduces the error probability as described above . it should be noted that other numbers of transmit - antennas , i . e . more than two transmit - antennas at the transmitter , and receivers , i . e . more than one receive - antenna at the receiver , could equally be used .