fig3 shows the block diagram of components of a first embodiment of the present invention using fm - cw as a means of determining distance in a pim measurement device . the fm sweep is introduced using sweep generator 40 . the sweep generator 40 is connected to f1 source 2 . for purpose of illustration , the sweep generator 40 is shown creating a 1 . 4844 us period saw tooth causing a +/− 11 . 25 mhz modulation ( ranging 22 . 5 mhz as shown ) that is added to f1 of source 2 . the fm sweep signal f1 and the fixed signal f2 when modified by the pim source will produce the additional signals 2 *( f1 + fm )− f2 and 2 * f2 −( f1 + fm ). the delayed in time ( distance ) signal 2 *( f1 + fm )− f2 will be mixed with an internally generated non delayed in time ( distance ) signal 2 *( f1 + fm )− f2 to produce the desired measurement signal δf . the δf signal represents 68597 × distance in feet . the distance to fault from p 1 can , thus , be determined by distance in feet = δf / 68597 the circuitry of fig3 has components added to accomplish addition of the sweep generator frequency and then downconversion of the output signal for analysis . to provide a signal source for downconversion , the circuitry includes a × 2 frequency multiplier 44 connected to the f1 source 1930 to provide a 3860 mhz output with a +/− 22 . 5 mhz sweep . the output of the × 2 frequency multiplier 44 is provided to a first input of a newly added mixer 42 . the second input of mixer 42 is provided from the f2 signal generator 4 , so that the output of mixer 42 provides a frequency 3680 − 2127 . 5 = 1732 . 5 mhz . bandpass filter 46 centered at 1732 . 5 mhz eliminates other mixing products from mixer 42 to provide a first input to downconverting mixer 48 . a second input of mixer 48 provides the reflected test signal at 1732 . 5 mhz from bandpass filter 32 . with both inputs of mixer 48 at 1732 . 5 mhz , they are downconverted at the output of mixer 48 , leaving substantially only the frequency deviation δf due to reflection of a test signal created by the pim source 30 . a lowpass filter 49 removes mixing products higher than 50 mhz , leaving only the forward trace sweep δf and retrace sweep signal 45 mhz − δf . measurement of δf with a digital receiver using fourier transform to time domain or a spectrum analyzer gives a measurement of total distance from port p 1 to the pim source using the formula distance in feet = δf / 68597 , with post processing to account for the both the forward sweep δf and retrace 45 mhz − δf . exemplary signal frequencies are shown in fig3 , as well as subsequent drawings , to illustrate operation of the circuitry and are not intended to limit the operation frequencies that can be used with the circuitry . further in fig3 and in subsequent drawings included with the specific frequency number a signal type indicator . as shown , a 1 unit thickness indicator represents a non - modulated signal , a 2 unit thickness indicator represents a modulated signal , and a 4 unit thickness indicator indicates a signal that is both modulated and doubled in frequency . also illustrated in fig3 are the forward sweep δf signal lobes and the retrace signal ( 45 − δf ) signal lobe illustrating the difference in these signals as generated by the circuitry of fig3 . fig4 shows an alternative to the circuitry to fig3 for determining distance to fault that will eliminate the retrace ( 45 mhz − δf ) from the output frequency . the result will be a detected frequency of δf = distance × 68597 for both the forward sweep and the retrace sweep . this eases the post processing burden as well as increases the signal to noise ratio of the measurement . the circuit of fig4 includes an additional oscillator 50 operating at f2 minus the sweep frequency of sweep generator 40 ( 2127 . 5 −= 2105 mhz in the example shown ). the output of oscillator 50 is applied to the second input of mixer 42 instead of the output of the f2 oscillator 4 that is connected in fig3 . thus , the output of the mixer 42 is now a signal at frequency 3860 − 2105 = 1755 mhz . the bandpass filter 46 , thus , has a center frequency adjusted to 1755 mhz . the output of bandpass filter 46 at 1755 mhz mixes in mixer 48 with the output of bandpass filter 32 to provide a signal at δf − 22 . 5 mhz for the forward sweep and 45 mhz − δf − 22 . 5 mhz = 22 . 5 mhz − δf for the retrace . these signals when mixed in mixer 51 with a fixed 22 . 5 mhz signal from oscillator 52 and provided through filter 53 will produce δf − 22 . 5 mhz + 22 . 5 mhz = δf for the forward sweep and 22 . 5 mhz − δf − 22 . 5 mhz =− δf for the retrace . the distance to fault from the port p 1 can , thus , be measured using the formula distance in feet = δf / 68597 without significant post processing . fig5 shows a block diagram of a pim measurement device using cw signals , but using a change in phase , as opposed to frequency shift as in fig3 - 4 , as a means of determining distance to fault . the sweep generator 40 uses a step frequency of 674 khz over 33 . 4 steps causing a +/− 11 . 25 mhz frequency change on f1 . the reflected pim signal will produce a phase change at phase detector 56 equal to distance in feet = δφ degrees / 1 . 097 the circuit of fig5 substitutes a phase detector 56 for the mixer 48 and lowpass filter 49 of fig3 . the inputs of the phase detector are provided from the outputs of bandpass filters 32 and 46 . the filters 32 and 46 provide equal frequency outputs , both shown as 1732 . 5 mhz , so that the phase detector 56 then provides the phase change measurement signal δφ . distance in feet from the port p 1 to the pim source can be determined in feet using the formula distance in feet = δφ degrees / 1 . 097 . no amplitude measurement of the pim signal , however , will be available from the output of phase detector 56 . fig6 shows modification to the block diagram in fig5 with a linear quadrature detector circuitry added allowing both amplitude and phase measurements of the pim signal . the circuitry includes a 0 / 90 degree phase splitter 64 with both a 0 degree and a 90 degree phase shifted output to provide imaginary and real signal components to first inputs of mixers 60 and 62 . the phase splitter 64 receives an input from bandpass filter 46 . a 0 degree power splitter 58 receives the output from bandpass filter 32 and provides second inputs to the two mixers 60 and 62 . the output of the mixer 60 is provided through low pass filter ( lpf ) 66 to provide an imaginary signal ( i ). the output of mixer 62 is provided through low pass filter ( lpf ) 68 to provide a real signal ( r ). as indicated in the fig6 the value for δφ = arctangent ( i / r ) with i being the imaginary component from mixer 60 and r being the real component signal from mixer 68 . the value of the pim magnitude can be calculated by a = squareroot ( i 2 + r 2 ). fig7 shows modification to the block diagram of fig5 that downconverts the varying frequency pim signals to fixed 200 khz offset baseband signals to allow measurement of distance to a pim source as well as magnitude of the pim source using low frequency a / d conversion . in fig7 an oscillator 70 is added operating at the frequency 2 ( f1 + fm )− f2 plus the desired fixed offset intermediate frequency ( fif ) to serve as a local oscillator for downconversion to baseband . here with the desired offset of fif being 200 khz , f1 being 1930 mhz and f2 being 2127 . 5 mhz , the frequency of oscillator 70 is shown at 1732 . 7 mhz . a connection from step sweep generator 40 is made to oscillator 70 to provide the sweep offset fm . also added in place of the phase detector 56 of fig5 are 0 degree splitter 58 mixers 60 and 62 and output bandpass filters 66 and 68 . in the circuitry of fig7 , the mixer 60 has an input receiving the reflected signal from the pim source through bandpass filter 32 . the mixer 62 has an input receiving a reference signal from the output of bandpass filter 46 . a second input of the mixers 60 and 62 receive the output of local oscillator 70 as provided through splitter 58 . the output of filter 68 , thus provides a reference signal downconverted to 200 khz , while the output of the filter 66 provides the reflected signal for measurement downconverted to 200 khz . the fixed 200 khz signals from filters 66 and 68 can be processed to measure pim amplitude as well as distance to fault using phase angle . fig8 shows the pim measurement circuit of fig7 with calibration components added to enable calibrating the distance , phase and amplitude for a one port pim reflection measurement . a non pim producing load 81 provides a first calibration component used to provide a phase and amplitude reference to measure the system &# 39 ; s inherent pim for cancellation . a known pim standard with a series load 82 is connected and measured relative to the standard load 81 as a second calibration component . finally , a pim source with a ¼ wavelength section of line length ( l ) at the frequency of 2f1 − f2 is attached as a third standard 83 , enabling the different length line to produce a 180 degree phase shift on the returned pim measurement for comparison . the resulting calibration will yield a full scale measurement of the known pim standards down to the system noise floor . fig9 is a block diagram showing modification of the circuit of fig7 to make a through pim measurement . the circuit includes both a port p 1 and port p 2 . the return path through bandpass filter 32 is disconnected from its connection to port p 1 in fig8 , and connected through a duplexer 84 to port p 2 in fig9 . an unknown pim source with cable 86 is connected between ports p 1 and p 2 for the through measurement . as illustrated , the two signals f1 = 1930 mhz and f2 = 2127 . 5 mhz are provided to port p 1 , while those signals along with signals of 2f1 − f2 = 1732 mhz and 2f1 + f2 = 2325 mhz generated by the pim source proceeds through port p 2 and duplexer 84 . the duplexer 84 sends the signals at 1930 mhz and 2127 . 5 mhz to a load , while the remaining signals are provided through bandpass filter 32 centered at 1732 . 5 mhz . thus , the 1732 . 5 mhz forward unknown pim signal is provided to mixer 62 , while the 1732 . 5 frequency reference signal is provided through mixer 60 for downconversion and subsequent measurement . fig1 shows modification to the measurement circuit of fig9 that allows for a through calibration of a known pim signal . the circuit of fig1 modifies the circuitry of fig9 to include an internal calibration switch 90 , a 0 degree splitter 91 a precision attenuator 93 and duplexer 92 . the calibration switch 90 allows connection of the precision calibrated 1732 . 5 mhz output of filter 46 to duplexer 92 when calibration is desired . the 0 degree splitter 91 enables the signal from filter 46 to be provided to the calibration switch 90 as well as to the mixer 62 for downconversion to provide a reference signal . the duplexer 92 connects to the calibration switch 90 and allows forward transmission of the reference pim signal level through switch 90 to port p 1 , as well as the signal from combiner 14 to pass to test port p 1 . the duplexer 92 prevents any signals other than the 2 *( f1 + fm )− f reference signal and f1 + fm and f2 from leaving port p 1 while preventing f1 + fm and f2 from entering the switch 90 . the thru line 94 is used during calibration along with calibration switch 90 so that the calibration signal from attenuator 93 is provided through test ports p 1 and p 2 and is then downconverted to a thru signal using mixer 60 . the thru signal is compared with the reference signal output from mixer 62 to enable the thru calibration . fig1 shows a full implementation of embodiments of the present invention as shown in fig7 through fig1 . the circuit of fig1 modifies the circuitry of fig1 to substitute calibration switch 90 with switches 102 and 104 . switch 102 provides the same function of switch 90 to connect or disconnect the output of 0 degree splitter 91 and attenuator 93 to duplexer 92 during through calibration . switch 102 in combination with switch 104 further allows the test port p 1 to be connected to the mixer 60 to provide a downconverted reflected signal for comparison to the reference signal measurement from mixer 62 . the switch 104 alternatively allows the test port p 2 to be connected to mixer 60 to provide a downconverted through ( t ) measurement , or to allow the switch 102 to connect test port p 1 to mixer 60 . the calibration components as well as pim measurement devices are combined as a pim calibration kit 106 and can be connected for test measurements as described with respect to fig8 through 10 . fig1 further shows that the frequency domain can be converted to time or distance calculation using a dsp 110 . the dsp 110 can be simply programmed to perform the distance calculation from measurements taken using the test setup of fig1 . the dsp 110 for determining distance can be a single device that also does processing of the , reflected and through signals for a test measurement . although shown as a dsp 110 , it is understood that a microprocessor , field programmable gate array ( fpga ), or a general application specific integrated circuit ( asic ) can be used to provide frequency to time domain conversion to determine distance to a pim either together or separate from other measurement circuitry . the dsp 110 is provided in combination with a dual a / d converter 108 . the dual converter 108 converts the analog forward signal from one analog port to digital , and converts the analog reverse signal or through signal at the other port to a digital signal for processing in the dsp 110 . although shown with a dual a / d converter 108 , separate simultaneously triggered converters can be used . as indicated previously , the system of fig1 provides pim measurements and differs from a standard vna by separating forward and reverse signals using duplexer 92 rather than a directional coupler . in one illustrative example , a pcs device is used along with an aws device to illustrate how frequencies can create passive intermodulation ( pim ). in the example , the pcs transmit operation frequency is 1930 mhz and the aws transmit frequency is 2127 . 5 mhz . the aws receive channel operates in the range of 1710 mhz to 1755 mhz . a δf of +/− 11 . 25 mhz centered at the pcs frequency of 1930 will produce an intermodulation frequency at 2 * 1930 − 2127 . 5 or 1732 . 5 mhz with a δf of +/− 22 . 5 mhz due to the 2 × multiplier in the intermodulation equation , or 1732 . 5 +/− 22 . 5 mhz . this will interfere with the aws receive channel range of 1710 - 1755 mhz , and the distance to pim measurement will indicate the amplitude of the interference as well as the distance to any defective component causing the pim . the minimum distance that can be measured to a fault in a distance to pim causing defect according to embodiments of the present invention can be easily calculated using the formula below . the minimum distance is related to the velocity of propagation for the cable . for a typical low loss cable having a propagation of 0 . 9 and using a sweep frequency fswp of 45 mhz , due to the 2f1 doubling of sweep frequency , can be calculated as follows : determination of distance from frequency caused by delay δf can similarly be easily calculated as described below with reference to fig1 . fig1 provides a graph of a stimulus signal and a delayed response signal caused by a pim reflection . the graph of fig1 plots change of frequency versus change in time . the difference frequency caused by delay δf shows the frequency change from the stimulus to the delayed response signal . similarly , the change of time δt shows a time delay between the stimulus and the response . an initial time ta and initial frequency fa are shown for the plots . further a frequency fb is shown illustrating the maximum frequency difference between the minimum frequency fa and maximum frequency of both the stimulus and delayed response signal . with reference to fig1 , the formula for difference frequency δf is as follows : for an fm cw linear saw tooth ramp , and assuming a non - aliased cable to pim distance of 100 m or 328 ft and vp of 0 . 9 , sweep time , tswp , is as follows : providing this value for tswp into the formula for δf and using the sweep frequency fswp of 45 mhz and vp of 0 . 9 , we obtain δf as follows : although the present invention has been described above with particularity , this was merely to teach one of ordinary skill in the art how to make and use the invention . many additional modifications will fall within the scope of the invention , as that scope is defined by the following claims .