referring to fig1 a supplemental inflatable restraint ( sir ) control for inflating an air bag ( not shown ) includes an accelerometer 10 which senses vehicle acceleration coupled to a microprocessor 12 which interprets acceleration data and decides when to deploy the air bags . this portion of the system is considered to be the deployment circuit . a deployment command from the microprocessor is sent to a firing circuit 14 which supplies ac pulses to one end of a firing loop 16 . the firing loop is grounded at the other end and comprises a vehicle wiring harness external to the firing circuit 14 and includes a firing element also called a squib or initiator 18 , and a capacitor 20 . the capacitor is small enough to limit a pulse due to shorting of the wiring harness to a current insufficient to deploy the firing element , but large enough for several cumulative pulses to cause deployment . the firing loop also has a substantial inherent inductance 22 which may range from 1 to 30 μh , for example , depending on the length and routing of the harness . a power supply 24 provides a reserve voltage v res which may be 15 or 25 volts , for example , for supplying the current to the firing loop 16 . a high side fet or charge switch 26 between the power supply 24 and the firing circuit admits current pulses to charge the capacitor 20 and a low side fet or discharge switch 28 couples a grounded return path 30 to the firing loop to discharge the capacitor . a high side gate driver circuit 32 controls the switch 26 and a low side gate driver circuit 34 controls the switch 28 . a current feedback arrangement triggers the switches such that when a deploy command is given the gate drivers are alternately enabled so that alternate current pulses flow in the firing loop . a first current sensor 36 comprises a small current sensing resistor 38 in series with the switch 26 and the power supply 24 , a reference resistor 40 in series with the power supply and a current sink 42 , and a comparator 44 having its negative input at the low voltage side of the sensing resistor 38 and its positive input at the low voltage side of the reference resistor 40 . the voltage across the reference resistor 40 is about 0 . 5 v , for example and the current sensing resistor 38 is selected to develop that same voltage at the desired switching current threshold . when the switch 26 is closed a current pulse will flow as determined by the impedance of the firing loop 16 . initially the comparator 44 output will be low but as the pulse current increases to the switching threshold the voltage decreases on the negative input and the comparator switches to yield a high output . as the pulse current passes its peak and then crosses the desired switching threshold , the comparator will switch back to a low output . an npn transistor 46 with a grounded emitter has its collector connected to v cc via a pull - up resistor 48 and its base is coupled to the comparator output . the transistor inverts the comparator output to produce a clock signal on clock line 50 which is normally high and then goes low after the beginning of a pulse and then goes high as the pulse diminishes and then crosses the switching threshold . a second current sensor 52 comprises a current sensing resistor 54 in the return line 30 , a comparator 56 having its positive input connected to the high side of the resistor 54 , a current source 58 and a reference resistor 60 serially connected between v cc and ground with the high side of the resistor connected to the negative input of the comparator 56 . this current sensor 52 operates much like the sensor 36 , producing a high output for high values of the discharge current in the return line and going low as the diminishing current pulse crosses the current switching threshold . the resistors are selected to yield the same current switching threshold as in sensor 36 . an npn transistor 62 controlled by the comparator 56 output connects the clock line to ground when the discharge pulse is high and effects a high clock signal when the switching threshold is crossed . the two transistors 46 and 62 form a clock circuit for producing a rising clock signal on line 50 each time a current pulse of either polarity flowing through the firing loop declines to the current threshold . these two transistors combined with a d type flip - flop 64 comprise a toggle circuit . the flip - flop has its clock input connected to the clock line 50 , and its / q output coupled to the data input . the q output is coupled through an and gate 66 to the high side driver 32 and the / q output is connected through an and gate 68 to the low side driver 34 . the deploy command signal from the microprocessor 12 is connected to each and gate to enable the gates . both outputs of the flip - flop 64 will change state at each rising edge of the clock signal thereby toggling the drivers 32 and 34 , if the deploy command is present , causing the switches 26 and 28 to simultaneously switch on and off in opposite phase . an intermediate value bias voltage v bias is applied to the firing loop primarily for diagnostic purposes . this has the effect of normally keeping the capacitor 20 partially charged . then when the microprocessor 12 first issues a deployment command and if the flip - flop / q output happens to be initially high , discharge current will flow in the return circuit 30 to trigger the current sensor 52 . if the q output is initially high the charge current will trigger the current sensor 36 . in either case the series of alternate polarity pulses will commence . the resulting firing loop current is shown in fig2 and 3 . in fig2 the circuit parameters are v res = 15v , capacitor 20 is 0 . 68 μf , loop resistance is 5 . 25 ohms and the inductance 22 is 1 μh . the switching threshold of the current sensors is set to 0 . 4 a . the parameters are the same for fig3 except that the inductance is 30 μh . it is obvious that the circuit dynamics altered the pulse frequency in response to the inductance values , but the rms current remained the same at 1 . 28 a as shown by the dashed line . the effect of the switching at the 0 . 4 a threshold ( both positive and negative ) is best observed in fig2 as a break in the trailing side of each pulse at the 0 . 4 a level . the same effect is shown in fig3 but is less evident . it is necessary to choose a switching threshold which yields an rms current which is sufficient ( higher than 1 . 2 a ) and which is substantially independent of the circuit inductance . this is possible as shown in fig4 which is a graph of rms output vs . switching threshold for various inductances . the outputs are all nearly the same for a switching threshold of 0 . 4 a which yields 1 . 28 a rms current at a supply voltage of 15v . other threshold values will result in various output currents and in some cases are below the level required for deployment . fig5 shows the rms output current for various values of loop inductance for this variable frequency circuit and for the prior art fixed frequency circuit operating at 100 khz . the fixed frequency control has an unacceptably low output for inductance values above about 7 μh whereas the variable frequency control technique has the same high output across the inductance range . it is apparent then that the expedient of current feedback for controlling the current pulses in a sir firing circuit permits a specific circuit design which operates properly over a wide range of loop inductance without special tuning to match to the load .