reference is now made to fig2 . this illustrates an embodiment of the invention which is similar in structure to that of the prior art as illustrated in fig1 . to simplify the description herein , like components in fig1 and 2 are identified with like character references with only the differences in the embodiment of fig2 being described in detail below . in the embodiment of fig2 , a splitter 30 is employed to split off a fractional portion 34 of the analog fm signal from the source 14 with the remainder portion 32 being supplied to the main transmitter 10 . the fractional portion 34 , which may be on the order of 1 - 10 % of that obtained from source 14 , is supplied by way of an adjustable phase adjuster 36 to a summer 40 , the opposite of splitter 30 . the phase adjusted fractional portion 34 from source 14 and digital iboc signal obtained from source 16 are combined by summer 40 and the combined signal , as a first combined signal , is supplied to the digital transmitter 12 . the fractional portion of the fm signal has been phase adjusted so that it will properly match that generated by the fm transmitter 10 when combined at the output combiner c . the outputs ν 2 and ν 3 are supplied to the coupler c and are combined to provide a combined or composite output ν 1 for application to the broadcasting antenna 20 . it has been determined that the new and improved embodiment in fig2 which may be considered as a dual fm ( dfm ) version , reduces the rf losses compared to that of the system in fig1 , making better use of the auxiliary transmitter 12 . there is also a reduction of the fm odd mode excitation that is achieved by feeding the in - phase power into the coupler . a lower fm odd mode excitation brings about various benefits including a lower overall fm insertion loss , and a tighter coupler coefficient on the order of 6 db or lower . the tighter coupling coefficient reduces the overall iboc power generation and this reduces the overall iboc power loss , enabling the injection of the fm signal through the auxiliary digital transmitter 12 . fig2 presents a dual fm injection system in that it provides high - level combination of the fm and digital signals . thus , the digital transmitter 12 carries the iboc signal and a fractional portion of the fm signal that has been phased so as to match and add at the output combiner c with the signal generated by the fm transmitter 10 . this system as shown in fig2 provides a solution of upgrading to iboc without impacting the output requirement of the main fm transmitter . this reduces by at least 50 % the power lost in the reject load . these features , along with the increase in overall system efficiency is described in greater detail in the analysis that follows below . the combining analysis hereinafter involves significant equations and their explanation with respect to various graphical terms . it is believed that various definitions of terms and language employed should be defined to facilitate an understanding herein . table 1 , presented below , provides these definitions . however , for incoherent signals , as those at different frequencies ( fm and iboc ), the total power is obtained by the sum of each coupled power . this can be proven by solving the square of the resultant instantaneous voltage . s 12 2 ν 2 ( t ) 2 + 2s 12 s 13 ν 2 ( t ) ν 3 ( t )+ s 13 2 ν 3 ( t ) 2 due to orthogonality between ν 2 ( t ) and ν 3 ( t ), their multiplication averages over time to zero , amounting to no contribution to the average power at the output . the resulting term , to be averaged over time , has been reduced to : which means that the average power at the output , for non - coherent signals , is obtained by summing each coupled input power , important to note that this value is smaller than the value obtained by the square of the sum of their rms voltages ( properly phased ), and accordingly , some power is being lost . in the following sections , the advantages are noted of adding some coherence between the signals ν 2 and ν 3 , such that the mid term does not totally cancel . partial coherence is obtained by adding some fm signal at port 3 , along with iboc . let &# 39 ; s now determine the minimum amount of dual fm injection that is needed so that the main fm power can remain unchanged . assume that ν 1 is the output voltage , ν 2 is the fm voltage applied at the thru port ( where the main fm transmitter connects ), and ν 3 — fm is the fm voltage applied at the coupled port of the output combiner ( where the digital transmitter connects ). to further simplify the analysis , make z o = 1 , and the total coupled fm power into port 1 ( output )= 1 . then at 100 % tpo , p 1 = 1 and ν 1 = 1 : p 1 =( s 12 ν 2 + s 13 ν 3 — fm ) 2 = 1 solving for ν 3 — fm , the fm voltage needed at the coupled port 3 to achieve output power p 1 = 1 : this expression is used for the condition of no impact on the main fm power . this condition forces ν 2 = 1 , and then the value ν 3 — fm obtained is the fm voltage needed at port 3 that guarantees that the main fm transmitter power is not affected . in other ways , viewed from the main transmitter &# 39 ; s port ( port 2 ), the combiner has no loss . if we now sweep a range of coupling factors , then we obtain the dual fm injection level that is needed , as a function of the coupler used , in order to make the main fm path lossless . so far it is seen that the required amount of dual fm injection at port 3 increases with the coupling factor , but in general it is a small fraction of the main fm service ( or tpo ). on the other hand , the required generation of iboc at port 3 is inversely proportional with the coupling coefficient . the smaller the coupling coefficient ( the looser the coupling ) the more iboc power will need to be generated . considering both effects at once , tighter coupling coefficients require more dfm injection but less iboc generation . the optimum coupling value that minimizes the size of the digital transmitter is now considered . this value includes fast , highly compressed envelope peaks . it is the short duration of these peaks , much shorter than the amplifier thermal constant ( its temperature will hardly change with modulation ) that allows the device to reach higher levels than in a cw mode . accordingly , in order to compare the size of a digital transmitter , whose envelope is being pulsed , to the corresponding 100 % tpo ( that is provided by a cw - operated fm transmitter ), we need to adjust these peak levels to its equivalent maximum cw operation . based on peak power measurements , we rate the maximum cw operation of a digital transmitter 0 . 5 db below its maximum pulsed level . as an example , a 11 . 22 kw peak - capable digital transmitter will be considered to be a 10kw cw unit ( 10kw = 11 . 22 kw − 0 . 5 db ). this is the value that will be used to compare sizes between the fm and digital transmitters . the relative size of the digital transmitter , respect 100 % tpo , can then be estimated by : if we sweep a range of coupling coefficients , and graph three different values of ν 2 ( main fm output ), we then obtain the results as seen in fig3 . several important conclusions can be extracted from fig3 . the optimum coupling coefficient ( for size ) is in the region of 5 to 6 db , instead of the 10 db coupling used in separate amplification . the digital transmitter needs to be 35 % the size of the main transmitter &# 39 ; s ( or tpo ), in order to provide the secondary fm injection level that is required to make the main fm - path lossless . it is important to note that this value ( 35 %) is comparable to the size demanded for separate amplification . in other words , with the same hardware ( as the one used in prior art ), we are now providing a “ lossless ” solution for the fm path , resolving the headroom problem . fig3 also shows that a bigger transmitter can afford to over - inject fm , resulting in gain at the main fm port . a smaller transmitter , with less injection than the one required ( not to impact the main fm path ), will make the main fm to have to overcome some loss . finally , by using now this tighter coupling coefficient ( 6 db ), the amount of iboc that needs to be generated drops by 4 db , of which 25 % will get coupled to the output . as a result , the dissipated iboc power into the reject load is about 30 % of the amount dissipated in separate amplification . the next section studies the reject power in more detail . p reject — fm =( s 13 ν 2 − s 12 ν 3 — fm ) 2 to keep our analysis practical , let &# 39 ; s select the same size transmitter as the one required for separate amplification ( minimum size ). with this restriction , the total fm voltage supplied at port 3 will be limited to : as the coupling coefficient is changed ( or swept ), and the digital transmitter size is now allowed to change , we will have to adjust the output of the main fm transmitter ( ν 2 ) in order to guarantee 100 % tpo at the output of the combiner ( ν 1 = 1 ) 5 . the value of the voltage at port 2 then needs to be : fig4 shows the total reject power for the dfm system , and its fm and iboc contributions . as a comparison , the straight line represents the reject power for a 10 db - coupler separate amplification system . the minima in total reject power is found to be somewhere between 6 to 7 db coupling , with a value equal to half of the power dissipated in separate amplification . considering signal clipping to be our limit ( independent of their probability of occurrence ), let &# 39 ; s calculate the maximum average power at which a digital transmitter can be operated , for different iboc injection levels . the total output power of the digital transmitter , relative to its cw equivalent rating , has been plotted in fig5 as a function of the iboc / fm ratio . the new operating average power , with a dfm injection of − 2 . 5 dbc , is about the same than the iboc - only rating . we can already predict an improvement in the overall system efficiency , i . e ., with the same generated average power of the digital transmitter ( respect separate amplification ), we have upgraded our system to fm iboc without demanding 11 % more power out of the main fm transmitter . the measured efficiency of a digital transmitter , operating at different levels of dfm injection is represented in fig6 . for iboc / fm ratios weaker than − 10 dbc , the efficiency of a digital transmitter is very similar to the efficiency of common amplification . for stronger injection levels , the efficiency quickly improves ( for the same average power ) and approaches the efficiency of iboc - only operation . knowing that the output power ( including iboc ) has to be 1 . 01 , the overall system efficiency is : we will assume a 60 % efficiency for the fm transmitter η 2 . the efficiency curve fit function used to generate the graph in fig6 will be used to compute the value of η 3 . equations for the other terms have already been determined . a graph of the overall efficiency for a dfm system implemented using the minimum size digital transmitter , is represented in fig7 . with the new dfm implementation , the efficiency approaches 50 %, as compared to separate amplification , whose efficiency would be 46 %. the coupling coefficient for peak efficiency is found at 6 . 75 db . we have described the dual fm injection , an improved high - level combining method of upgrading to iboc . this method uses a tighter coupling coefficient at the output combiner , which decreases the iboc losses . simultaneously , by injecting a small fraction of the fm signal , properly phased , at the coupled port , the total combining efficiency is increased , as both the main and the dual fm injection are partially summed at the output . a coupling factor of 6 db is the optimum value to minimize the size required for the digital transmitter and to reduce the dissipation in the reject load , and as such boosting the overall system efficiency by 3 to 4 %. under these conditions , the total reject power is expected to be reduced by 50 %. all these benefits are added to the fact that with dfm injection , the main fm transmitter does not have to increase its output level to overcome the coupler losses , an inconvenience at times unsurpassable for many installations in the field . finally , the system performance is dependent on the size of the digital transmitter in the following manner ; the bigger the digital transmitter to be used , the stronger the fm injection through it and the tighter the optimum coupling coefficient can be , this way obtaining a lower output power demanded of the main fm transmitter , a lower reject power and a higher overall system efficiency . it has been determined that this system of high - level combining with dual fm injection achieves lower insertion loss for the fm transmitter . instead of a 10 % insertion loss known in the prior art , the typical insertion loss for the new system is 0 db ( no loss ), if a 6 db coupler is used . if tighter couplers are used , an actual gain is achieved , as more fm power is added through port 2 . this improvement is fundamental to economically upgrade to iboc in “ difficult powers ” where the fm insertion loss can not be tolerated if prior art is to be used . overall efficiency of this new system can be as high as 52 % using present broadcast technology , which results in a reduction of 20 % in dissipated power . the rf combining efficiency can reach 98 . 5 % with typical values of 95 %. the equipment ratio to make the conversion is now 1 . 35 . the small difference of 4 % in size ( minimum ), in practice is enough to provide optimum configuration to broadcast customers . although the invention has been described in conjunction with a preferred embodiment , it is to be appreciated that various modifications may be made without departing from the spirit and scope of the invention as defined by the appended claims .