embodiments of a method for transmitting signals using antenna diversity , for instance in mobile communication systems , transmitter , receiver and computer program product therefor are described herein . in the following description , numerous specific details are given to provide a thorough understanding of embodiments . one skilled in the relevant art will recognize , however , that the invention can be practiced without one or more of the specific details , or with other methods , components , materials , etc . in other instances , well - known structures , materials , or operations are not shown or described in detail to avoid obscuring aspects of the invention . reference throughout this specification to “ one embodiment ” or “ an embodiment ” means that a particular feature , structure , or characteristic described in connection with the embodiment is included in at least one embodiment . thus , the appearances of the phrases “ in one embodiment ” or “ in an embodiment ” in various places throughout this specification are not necessarily all referring to the same embodiment . furthermore , the particular features , structures , or characteristics may be combined in any suitable manner in one or more embodiments . an embodiment of the solution provided herein refers to the case of the use of four transmission antennas designated respectively by tx 1 , tx 2 and tx 3 , tx 4 . the solution described herein can , however , be extended also to a larger number of antennas . this can be obtained in a simple way both by varying the length of the interleaving on the additional pairs of antennas and by using another channelling / spreading code for these antennas , albeit maintaining unvaried the data rate . the implementation of such extensions on the basis of what is described herein constitutes for a person skilled in the art a design task such as not to require a further detailed description herein . in this connection , it is once again to be noted that the aforesaid extensions do not in general entail an added burden in terms of hardware in so far as the generic base station of a third - generation mobile communication system ( bts 3 g ) must already be able to transmit all of the codes simultaneously . in the diagram of fig4 , it is assumed that there is at input a flow of data s 1 , s 2 , s 3 , s 4 coming from a generic modulator a known type ( m - psk or m - qam ). said flow of data is sent to a block 10 capable of performing simultaneously a demultiplexing function ( dmux ), together with a permutation function . basically , the module 10 splits the flow of input data between two lines designated , respectively , by 12 and 14 . on the output line 12 , the data flow is sent without undergoing any variation , then to be transmitted to an sttd encoder 16 built in accordance with release &# 39 ; 99 of the 3gpp standard . there are then provided subsequent spreading operations with a code c 1 implemented in blocks 181 , 182 in view of forwarding to the antennas tx 1 and tx 2 after prior combination , in two adder nodes designated , respectively , by 201 and 202 , with the respective pilot flows , which are to be used by the receiver for channel estimation . the portion of the transmitter associated to the output line 14 of the block 10 is structurally similar , in the sense that this too comprises an sttd encoder designated by 22 , with associated thereto at output two spreading modules 241 , 242 , which are to generate signals with a correspondingly widened spectrum . these signals are then supplied to the antennas tx 3 and tx 4 after addition of the respective pilot flows in two nodes designated by 261 and 262 , respectively . the basic differences between the two “ channels ” coming under the lines 12 and 14 are the following : whereas on the line 12 there is present the unaltered flow of data , just as it comes from the modulator at input to the block 10 , on the line 14 there is present a data flow in which each set of four symbols is temporally swapped ( and consequently subjected to shuffling or interleaving ) by subsets , typically in pairs , causing the sequence ( s 1 , s 2 , s 3 , s 4 ) to become , at output from block 10 , the sequence ( s 3 , s 4 , s 1 , s 2 ); and the spreading operation performed in the blocks 241 and 242 uses a second code c 2 , different from the code c 1 used by the spreading blocks 181 , 182 ; in other words , the two pairs of antennas tx 1 , tx 2 , on the one hand , and tx 3 , tx 4 , on the other hand , use different spreading codes , i . e ., c 1 and c 2 , respectively . the corresponding reception and decoding system , illustrated in fig5 , contemplates the presence of a receiving antenna rx , which is to receive , in a combined way , the signals coming from the transmission antennas tx 1 , tx 2 , tx 3 and tx 4 . the signals received present , of course , the typical alterations induced by propagation in the transmission channel c , namely , the addition of noise n and the presence , in the signals received by the various transmission antennas , of phenomena of multipath fading that act in different ways in regard to each signal ( this fact being , precisely , at the basis of the operation of diversity techniques ). the signal coming from the receiving antenna rx is sent to two matched filters 301 and 302 , which are to perform the de - spreading operation , eliminating the two spreading codes c 1 and c 2 introduced in the transmission stage . the operation of the filters 301 , 302 is based upon the formulae given in what follows , which have been developed just for the case of just one antenna in reception but can be extended ( according to criteria that are evident to a person skilled in the art ) to the case of more than one antenna in reception . in particular , the signal received in four signalling intervals can be expressed in the following way : r 1 = s 1 h 1 ′ c 1 − s 2 * h 2 ′ c 1 + s 3 h 3 ′ c 2 − s 4 * c 2 r 2 = s 2 h 1 ′ c 1 − s 1 * h 2 ′ c 1 + s 4 h 3 ′ c 2 − s 3 * h 4 ′ c 2 r 3 = s 3 h 1 ″ c 1 − s 4 * h 2 ″ c 1 + s 1 h 3 ″ c 2 − s 2 * h 4 ″ c 2 r 4 = s 4 h 1 ″ c 1 − s 3 * h 2 ″ c 1 + s 2 h 3 ″ c 2 − s 1 * h 4 ″ c 2 − h 1 ″ and h i ″ represent the channel coefficients , and − c 1 and c 2 are the two spreading codes used on the two pairs of antennas tx 1 , tx 2 and tx 3 , tx 4 , respectively . in the example given above , there has been considered , for reasons of simplicity , just one path from the generic transmitting antenna to the receiving antenna , but the mathematical expression given above can be readily extended — as is evident for a person skilled in the art — to the case of propagation on multiple paths , in general on n different paths . the channel coefficients h 1 ′ and h i ″ are assumed as being more or less constant ( or estimated to be such ) on two symbol time intervals . the corresponding estimation , implemented according to what is proposed by the 3g standard in release &# 39 ; 99 ( but also already starting from release 5 ), is performed according to known criteria in a channel - estimation block 32 that sends the corresponding coefficients to a linear receiver 34 , which is to supply at output the symbols received s 1 , s 2 , s 3 , s 4 . it will , however , be appreciated that the solution described herein , as regards the channel - estimation function , is in no way tied down to the adoption of the specific technique described in the 3g standard . the solution described herein can in fact be used also together with other estimation methods . after executing the de - spreading operation , in the first two symbol time intervals , there is obtained : s 1 ′= h 1 ′* r 11 + h 2 ′ r 21 * s 3 ′= h 3 ′* r 12 + h 4 ′ r 22 * whence we obtain the estimates s l ′ s 2 ′ s 3 ′ s 4 ′. s 1 ′= h 3 ″* r 32 + h 4 ″ r 42 * s 2 ′=− h 4 ″ r 32 + h 3 ″*+ r 42 s 3 ′= h 1 ″* r 31 *+ h 2 ″ r 41 * s 4 ″=− h 2 ″ r 31 *+ h 1 ″*+ r 41 whence we obtain the estimates s 1 ″ s 2 ″ s 3 ″ s 4 ″, where finally , the estimates of the four symbols received are extracted by summing the two sets of partial estimates according to the relations : the estimates in question constitute , precisely , the output signals indicated in the diagram of fig5 by s 1 s 2 s 3 s 4 . the above convention has been adopted for reasons of simplicity , taking into account the fact that , clearly , the estimation of the output signals corresponds exactly to the input signals transmitted , i . e ., the signals sent at input to the block 10 of fig4 in the case of ideal operation of the system . it will be appreciated that the decoding technique just described can be readily extended to the case of m generic pairs of transmitting antennas with m & gt ; 2 . also in this case , the result can be obtained simply ( and according to criteria that are evident for a person skilled in the art on the basis of the indications here provided ) both by varying the interleaving length on the pairs of additional antennas , and by using another channelling / spreading code for them . also at the receiver end , it is possible to use a number of receiving antennas . given that the reception system is linear , using the same method just described for each receiving antenna , the total estimate of the symbol will now be given by the sum of the various contributions of estimation supplied by each receiving antenna . the tests conducted by the present applicant show that the adoption of an embodiment of the technique just described leads to considerable advantages in terms of performance . this applies as regards the performance in terms of bit - error rate ( ber ) and according to a direct comparison with the proposals currently under debate at the 3gpp ( usually indicated by the post - fix “ rel 5 ”). with respect to the known solution referred to above , one embodiment of the technique described herein moreover enables elimination from the base station of the phase - rotation function and the corresponding circuit ( both at a hardware level and at the level of software components ) by maintaining , at the same time , the so - called full rate ; in other words , space - time coding does not reduce the transmission data rate . the results of the comparisons to which reference has been made previously used the same channel - estimation system proposed by the standard . once again , the demodulation technique described herein is maintained in the linear form , involving just de - spreading on two or more codes , which can be rendered perfectly serial and in line with a possible adoption of multiple - code transmission for a single client already envisaged by the standard . of course , as already indicated previously , instead of using , as in the example embodiments illustrated herein , two different scrambling codes c 1 , c 2 , with a solution that is altogether equivalent it is possible to keep the same scrambling code for the two sets of spreading blocks 181 , 182 and 241 , 242 illustrated in fig2 , using , however , for the two channels corresponding to the lines 12 and 14 , at output from the module 10 , two different codes of the ovsf type ( or of any other type that can be used in a cdma scheme ). consequently , without prejudice to the principle of the invention , the details of implementation and the embodiments may vary widely with respect to what is described and illustrated herein , without thereby departing from the scope of the present invention , as this is defined in the claims that follow . all of the above u . s . patents , u . s . patent application - publications , u . s . patent applications , foreign patents , foreign patent applications and non - patent publications referred to in this specification and / or listed in the application data sheet , are incorporated herein by reference , in their entirety .