although the following detailed description contains many specifics for the purposes of illustration , anyone of ordinary skill in the art will appreciate that many variations and alterations to the following details are within the scope of the invention . accordingly , the following preferred embodiment of the invention is set forth without any loss of generality to , and without imposing limitations upon , the claimed invention . fig1 provides a general view of the system architecture of a base station according to the present invention . the base station comprises a receiving antenna array 10 having n antenna elements . in this embodiment , the system also comprises a separate antenna array 15 for transmission . using antenna duplexers , however , the arrays can be combined , as is well known in the art . the embodiment allows for low cost duplexers and antenna filters since much less power is required per element to provide the required effective radiated power ( erp ) due to beam forming . preferrably , the number n of antenna elements is approximately 16 . each of the n antenna elements is coupled to a corresponding one of a set of n conventional receivers 101 . each receiver down - converts an incoming signal in frequency and digitizes it to produce a received signal having i & amp ; q ( in phase and quadrature ) signal components . in this embodiment , the receivers are coherently tuned by a common local oscillator 104 to allow for both phase and amplitude data measurement , thereby producing , at any given instant , an n - dimensional received signal vector having complex valued components . alternatively , a calibration signal of fixed frequency can be injected to all receiver channels simultaneously with the received signal , allowing for continuous estimation of the phase and amplitude difference among the receivers . the calibration signal can be differentiated from the received signal since it is not spreaded and can have a very low level since its integration can be very long . specific relevant receiver designs are presented in u . s . pat . no . 5 , 309 , 474 . the received signal vector from the n receivers 101 is fed to a set of l channel estimators 11 and also to a corresponding set of l receiver banks 14 . each channel estimator 11 and corresponding receiver bank 14 is used to estimate the channel and receive the signal from a single mobile unit . thus the maximum number of mobiles that can simultaneously be served by the base station is l . in a preferred embodiment , l is at least 100 . the estimators 11 are identical to each other in both structure and principles of operation . similarly , the receiving banks 14 are also identical . accordingly , the following description is limited to a single estimator 11 and its corresponding receiver bank 14 which serve to estimate the channel of a single mobile unit and receive its signal . in the preferred embodiment , channel estimator 11 comprises a set of n despreaders 102 , a corresponding set of n fast hadamard transformers ( fhts ) and a spatial correlator 105 . the despreaders 102 are conventional code - correlators described in detail , for example , in u . s . pat . no . 5 , 309 , 474 . each of the n despreaders correlates a single component of the received signal vector with a pseudo - noise ( pn ) code sequence assigned to the associated mobile in accordance with the is - 95 cdma standard . each code correlator uses a variable time offset ( synchronized with the other code correlators in the same bank ) to separate multipath parts that arrive with at least one pn chip period difference . the following description discusses the processing of one multipath part . all multipath parts that are strong enough to be isolated are processed identically . each correlator 102 outputs a despread signal corresponding to one multipath part of the signal from one mobile received at one antenna . this despread signal is fed into a fast hadamard transformer ( fht ) 103 . the fht used in the present invention is identical to conventional fhts ( described , for example , in u . s . pat . no . 5 , 309 , 474 ), except that the fht of the present invention retains the complex phase information of the input . in other words , whereas the standard fht outputs are converted to magnitudes , the fht used in the present invention outputs complex numbers , thereby preserving both phase and amplitude data . each fht in this embodiment has 64 complex outputs , whose magnitudes represent the degree to which the despread signal correlates to each of the 64 symbols in a predetermined symbol alphabet . in the preferred embodiment , the symbol alphabet is a set of 64 orthogonal walsh symbols . for a given symbol received at the array 10 ( in is - 95 , a symbol period is approximately 205 microseconds ), the signals received at the n antenna elements are , separately and in parallel , passed through n respective receivers 101 , despreaders 102 , and transformers 103 , while retaining the relative phase information of the signals . the collection of n fhts together produce an n × 64 signal matrix b of complex elements . each column of b is an n - dimensional vector , called the spatial response vector , whose n components represent the correlation of one walsh symbol with the signal received at the n antenna elements . the matrix b is fed column - by - column to the spatial correlator 105 following timing synchronized to the walsh symbols . as will be described in detail below in reference to fig2 spatial correlator 105 correlates the signal matrix b with an array calibration matrix a to produce a correlation matrix c that represents the correlations of the signal received at the antenna array with both a set of predetermined directions and a set of predetermined symbols . from an analysis of the matrix c the correlator produces a signal angle of arrival ( aoa ), and a scalar value ( aoa quality ) that is proportional to the &# 34 ; purity &# 34 ; of the wave front and the signal level . this data is transferred to a controller 106 that uses it to determine the best uplink beam coefficients for this particular signal part . typically , this entire process is performed for the four strongest multipath parts . in addition , time of arrival ( toa ) and aoa certainty data are produced , allowing for the generation of a spatially matched filter that contains beamforming information for each signal part . the functions of the channel estimator 11 described above are performed in parallel in all the other channel estimators for the other mobiles being handled by the base station . the controller 106 receives beamforming information from each of the channel estimators 11 . thus controller 106 obtains spatial information regarding all the signal parts from all the mobiles . the controller 106 then downloads this information , in the form of coefficients , to the receiving banks 14 which use the spatial information from the channel estimators 11 to improve the reception of the signals from the mobiles . each receiving bank 14 comprises beam formers 112 to form narrow beams towards the signal parts associated with a single mobile . because the strong signal parts are selectively detected , the beam former creates a well matched spatial filter for the incoming signal , including its multipath components . the beamformers 112 feed spatially filtered signals to the four fingers of a conventional is - 95 rake receiver 113 ( described in u . s . pat . no . 5 , 309 , 474 ). it should be noted , however , that the beam former outputs can be fed to other receiver types . as a result of the spatial filtering process described above , the carrier - to - interference ( c / i ) ratio is significantly improved over conventional cdma systems . the improvement in c / i is about the ratio between the effective beamwidth created ( about 10 to 30 degrees ) to the existing antenna beams ( about 100 to 120 degrees ). note that the aoa and toa data are also transferred to the central controller 120 where the system determines the most optimal downlink beams configuration . the downlink process will be discussed later , as part of the explanation of fig4 . although the preferred embodiment uses an is - 95 based architecture , the above process can be implemented with any wireless protocol that makes use of a finite alphabet or training sequence . for example , in gsm systems a training sequence is available in every wireless burst . since the training sequence is known , a correlation between the incoming signal and a stored training sequence at the receiver will produce same results as described above ( provided frequency error is not too great relative to the sequence length ). the correlator 102 and the hadamard transformer 103 are replaced in this case by a training sequence correlator ( convolver ). since there is only one possibility for a training sequence , it is not required to try for many possibilities as done by the hadamard transformer in the preferred embodiment . fig2 illustrates the details of the spatial correlator 105 . in this embodiment the spatial correlator is a stand - alone unit . due to redundant functionality between this unit and the current implementation of is - 95 rake receiver , however , the spatial correlator can be integrated with the rake receiver . the preferred embodiment is optimized to is - 95 ( uplink utilizing m - ary modulation ), however , any signal that has either a finite alphabet ( limited number of symbols ) or a training sequence can utilize the same idea . the use of the known signal structure facilitates simple array response vector determination and eliminates the necessity for complex covariance matrix calculation and analysis . hence , this approach can be utilized for gsm and tdma wireless air - interfaces as well . the columns of the signal matrix b ( i . e . the spatial response vectors from the fhts ) pass through a mux 206 and are then correlated with the columns of the array calibration matrix a which is stored in a ram 203 . in abstract terms , the correlation process is performed by multiplying the conjugate - transpose of the calibration matrix a by the signal matrix b . the result is a correlation matrix c = a h b . it is important to note that this abstract calculation may be implemented in many different ways , all of which are mathematically equivalent to each other . the calibration matrix a is also known as the array manifold and is generated by measuring the antenna array response in antenna test range . each column of a represents the response of the antenna array in one of a predetermined set of directions . for example , if the angular space is divided into 360 directions , then each of the 360 columns of a is an n - dimensional vector representing the response of the n antenna array elements in a given direction from the array . in the computation of the matrix c , these 360 vectors are spatially correlated with the 64 columns of the signal matrix b to produce a 360 × 64 element matrix , where element i . j represents the correlation of the received signal with the j th symbol in the i th angular direction . in the preferred embodiment , the correlation is performed very efficiently through the use of a unique and simple calibration table representation which allows the matrix multiplication to be implemented without any multiplications . each complex valued entry of the calibration table matrix a is quantitized such that both real and imaginary parts are each represented by two bits only . more specifically , each part is represented by one numerical bit plus one sign bit , thus : ( 0 , 0 )=- 0 , ( 0 , 1 )=+ 0 , ( 1 , 0 )=- 1 , ( 1 , 1 )=+ 1 . each complex valued entry is therefore represented by just four bits . the reduced resolution in this simple quantization scheme is compensated by increasing the number of array elements to about twice relative to current base station arrays . this simple bit - plus - sign data structure allows the vector dot products between the matrix columns to be calculated using a complex adder 204 . in conventional implementations , the vector dot product would require a collection of n multipliers . the technique of the present invention , therefore , dramatically simplifies the implementation of the spatial correlation operation . a timing generator 201 synchronizes the spatial correlator process to the walsh symbol period ( i . e . the end of the hadamard transform ) that is derived from the base station pilot timing . the n × 64 signal matrix is latched into the mux circuit 206 which provides the column vectors to the complex adder 204 one at a time . for each vector , the complex adder 204 performs separate correlations of the vector to every one of the 360 columns of the calibration matrix . the because the calibration table data are only 0 , 1 , or - 1 , they are used in the complex adder to decide whether to null , add , or subtract each element in the vector . a ram address generator 202 is driven also by same timing generator 201 to synchronize the presentation of 360 columns of calibration data with each lached vector . note that the number of array elements n does not change the correlation matrix dimensions , which are determined only by the number of predetermined symbols in the alphabet and the number of predefined angular directions . the correlation matrix c is stored in a spatial correlation ram 207 and processed by a maximum value selector 205 that is a simple serial comperator in the preferred embodiment . the end result of the spatial correlator process is the best expected aoa for the selected signal part and the associated &# 34 ; inner product &# 34 ; value ( used as a certainty factor ). this result is reported to the controller 106 only if a preset threshold has been crossed . this threshold value is updated from time to time as necessary . when the threshold has been crossed , the controller registers the time offset associated as the signal part toa . this information is used to estimate the mobile unit range from the base station . it is possible to identify more than one maximum at a time utilizing a recursive process : after identifying the largest value in the correlation matrix , the neighboring matrix elements are ignored ( ignoring the neighboring elements minimizes the probability for &# 34 ; non peak &# 34 ; selection ) and another &# 34 ; peak &# 34 ; search is executed . this feature allows the identification of multipath parts that cannot be differentiated in time alone ( as done in existing rake receivers ), allowing for beam forming reception of small time spreaded multipath . this approach has great advantage for close to bts mobile unit communication . the threshold value is calculated by averaging the reported results over long averaging period &# 34 ; window &# 34 ;. for example , k reported results are accumulated at the controller and the accumulated result is divided by k . since most of the reported results are generated by non - time correlated elements , the results are &# 34 ; noise like &# 34 ; and averaging them provides a good estimate of the channel noise level . since the channel noise is a linear function of the number of active mobile units , this level needs to be updated from time to time as stated . fig3 details the uplink beam former 112 of fig1 . in this embodiment , the uplink beamformer is presented as a stand alone unit . however , it is possible to integrate the beam former into the channel estimator due to the &# 34 ; bit plus sign &# 34 ; arithmetic that makes it a very low gate count device . signal outputs from the n base station receivers 101 are fed into a complex adder 304 for beamforming . since the data rate for is - 95 is about 10 mega - samples per second , the complex adder 304 can execute at least 4 vector sums per one vector data sample using present technology . the beamforming coefficients are downloaded from the controller as described above into a coefficient ram 303 . a timing generator 301 and an address generator 302 cause the coefficients to &# 34 ; rotate &# 34 ; into the complex adder . the coefficients are used as described above in reference to the spatial correlator of fig2 to form a dot product using only complex addition . the vector summation result is fed into an interface unit 305 for transferring the result to the rake receiver modem . in other embodiments , any finite alphabet or training based protocol based modem could be used . the effect of the beamformer is to spatially filter the incoming signal to preferrentially select for signals arriving from the known directions of the signal parts of a particular mobile unit . signals from other directions are attenuated , and the reception of the desired signal is naturally improved . fig4 illustrates an example of a spatial distribution of downlink beams . downlink management is quite different from the uplink since is - 95 is not a symmetrical protocol and uplink frequency is different from the downlink frequency by at least 60 mhz ( cellular ). the difference in frequency causes the uplink and downlink channels to be non - correlated . the aoa & amp ; toa of the uplink and downlink , although statistically similar , may differ significantly . hence , the downlink can be only statistically estimated based on data collected in the uplink , as described above . in addition , the downlink requires broadcasting of a pilot signal to associated mobile units . as a result , individual downlink beams are not possible ; only &# 34 ; mobile group &# 34 ; beams are realizable . hence , the downlink approach is based on a combination of wide and narrow beams determined by data collected in the uplink . the beam configuration is determined by the mobile unit distribution around the base station . the wide beams are required to assure proper coverage at closed proximity to the base station where most of the downlink signal arrives to the mobile unit by way of scattering from nearby reflectors . the system adjusts the wide beams 401 to assure proper coverage for the mobile units close to the base station . the narrow beams 402 are adjusted mainly to accommodate &# 34 ; far away &# 34 ; mobile units . since most of the mobile units will be in the outer coverage area , the narrow beams are expected to service the majority of the mobile units . increasing the number of downlink beams causes the increase of softer handoff , thereby countering the increase in capacity . hence , assigning beams in the downlink must be done very carefully . we assume a uniform mobile unit distribution , and a maximum illumination of q , that is the maximum number of simultaneous transmission channels including softer handoff . the term q * p is the number of mobile units that come in with high angular spread , called &# 34 ; wide angles &# 34 ;. q * p / n is the portion of the portion of wide angles that are within the narrow beam , and are all in softer handoff , thus , adding to the illumination in the overlapped sector twice . if , as a result of the beams combination x mobile units can be added , x * p additional wide angle type are added ( assuming p remains as before ), out of , x * p / n are following the same rule as for the q * p / n above . in the narrow beam space , we get q ( 1 - p )/ n + x ( 1 - p )/ n mobile units , but due to some handoff caused by the overlap we must increase the value of their illumination by factor 1 + b . b can be kept small since outer cell associated mobile units will naturally prefer the narrow beams . fig5 is a graph of the capacity increase ratio with respect to both the number of narrow beams and the probability / 10 of wide angular spread multipath with softer handoff probability fixed at 20 %. fig6 is a graph of the capacity increase ratio as a function of handoff probability / 10 and the wide angular spread multipath probability / 10 for 4 narrow beams . fig7 is a graph of two cases of wide angular spread multipath for variable handoff ratio and 4 narrow beams . following the above analysis , the enhancement of capacity for 4 narrow beams within one wide beam is approximately two . if the mobile unit distribution is non uniform , the enhancement can be even higher . fig8 is a graph of the expected capacity ratio for different density variance of mobile units . this improvement requires the narrow beam borders to be adjusted to avoid mobile unit density peaks . this adjustment function must be gradual to avoid excessive handoff while changing the beams . fig9 presents the a flowchart of the downlink beamforming determination process . mobile unit spatial data is collected and stored in memory . this data is then used to evaluate the mobile unit distribution around the base station by sorting the data into a two dimensional histogram . the histogram &# 34 ; peaks &# 34 ; are identified as follows : a two dimensional &# 34 ; smoothing &# 34 ; filter is executed to eliminate noisy histogram &# 34 ; spikes &# 34 ; and a common two dimensional &# 34 ; peaks search &# 34 ; process is utilized . for a system that is capable of forming m downlink beams , m &# 34 ; peaks &# 34 ; are sorted . the following is a closed loop process that adjusts the boundaries of the downlink beams and equalizes the number of mobile units associated with them . narrowing the beams will cause some mobile units to handoff to different pilots leaving only mobile units close to the selected &# 34 ; peak &# 34 ; hanging on to the associated pilot . this process proceeds at a very slow pace to avoid excessive handoff . fig1 presents an aparatus to generate an antenna array manifold ( calibration table ). an antenna array 601 , incorporates a collection of antenna elements , installed on a support mast that is connected to a turn - table 504 . the controller 506 commands the turn - table to rotate in pre - determined angle steps . a network analyzer 505 transmits through a transmitting antenna 502 an rf signal which is received by the antenna array 601 . the signals received at the elements in the antenna array are routed through an rf switch 503 to the network analyzer for measurement . in the preferred embodiment , the antenna array is circular but the invention can be implemented with any arbitrary array shape . the rf signal collected for each antenna element in this case can be written as follows : ## equ2 ## v represents the array manifold function , k is the element number and θ is the relative angle of arrival ( created by rotating the array relative to the rf signal source ). m is the total number of antenna elements in the circular array . λ is the rf signal wavelength . the data is collected and stored in a data storage unit and controller 506 . fig1 presents an embodiment illustrating a training sequence convolver , which may be used instead of the despreader 102 and fht 103 in some wireless standards . a data register 702 is a first - in - first - out ( fifo ) unit with a word bandwidth that is matched to the receiver i & amp ; q output width . the i & amp ; q samples are shifted through the data register 702 in two &# 39 ; s complement format . xor gates are used to compare the most significant bit of i and the most significant bit of q with bits of a training sequence stored in a training sequence register 703 . the resultant xor output are fed to an adder 701 and used to determine whether to add or subtract each i & amp ; q sample in the data register . the output of the adder is updated for every sample cycle and compared against a threshold in a magnitude threshold detector 704 . when the threshold is exceeded , the i & amp ; q values are registered as a component of the signal response vector that is then sent to the spatial correlator explained above . fig1 presents an embodiment of the invention that includes both searching and tracking functions ( in angle & amp ; time ). the addition of angular tracking increases the ability of the system to efficiently direct the receiving beams at all times . a searcher 80 acquires new multipath parts as before while a tracker 83 tracks them . the principle of operation of this embodiment is very similar to the embodiment described in fig1 . the main difference relative to fig1 is the addition of the tracker 83 . the n receiver outputs are fed in parallel to beamformer 812 . the controller 81 downloads to the beam former 812 not just one , but two beamforming information sets for each signal part to be tracked . the two sets correspond to two adjacent columns in the calibration matrix . this allows the beam former to continuously &# 34 ; toggle &# 34 ; between two angularly adjacent beams . the beam former output is fed into an &# 34 ; early / late gate &# 34 ; module 813 known in the art . the result of the combined &# 34 ; toggling &# 34 ; beam former and the &# 34 ; early / late gate &# 34 ; is in the form of four level values corresponding to : left beam / early time , right beam / early time , left beam / late time and left beam / late time . since the tracker is designed to track four multipath parts simultaneously , the results are reported to the controller through a multiplexer 815 . the controller 81 directs the beamformer and the &# 34 ; early / late gate &# 34 ; to balance all the four values above the same level by exchanging the beam former coefficients and advancing / delaying the gate &# 39 ; s clock . angular tracking is achieved by equalizing the right and left associated results while the time tracking is achieved by equalizing the early late associated values . this embodiment assures sufficient integration for reliable tracking . the sets of coefficients are entirely replaced when the searcher finds a multipath part that generates a significant higher level outputs than the ones tracked . in this embodiment , each channel is assigned it &# 39 ; s own downlink beamformer 830 . note also that this embodiment supports an individual beam for each active channel . fig1 presents an overview of a base station that employs channel estimators / trackers / beamformers described in fig1 . the antenna array 90 is coupled to a set of receivers 901 which are all driven by common local oscillator 904 , as in fig1 . the receivers output is placed on a data bus 910 to feed a plurality of channel estimators / trackers / beamformers 905 , each of which provides a bts channel element 906 with a plurality of signal parts . element 906 can be a rake receiver / data transmitter of is - 95 . the channel elements are feeding downlink data to the channel estimator / tracker / beamformers , which feed beamformed data to summation unit 907 . the summation unit outputs summed beam formed data to the bts transmitters 909 that are driven by common local oscillator 908 . the transmitter outputs are radiated through transmitting antenna array 91 . the above embodiment of the downlink requires additional &# 34 ; pilots &# 34 ; when applied to a cdma is - 95 base station . this may require some changes in the network control and network pilots allocation design . the following embodiment alleviates this requirement by distributing the overhead channels ( pilot , paging and synch .) through a wide beam while the traffic channels are individually transmitted through narrow beams directed at the associated mobile units . this approach does not change the conventional bts softer handoff profile , hence it does not require any changes in the network architecture . this proposed arrangement is facilitated by careful array beam synthesis techniques that are well known in the art . in particular , the beams are constructed to be phase matched in the mobile unit &# 39 ; s scattering region . the beams &# 39 ; coefficients are calculated to achieve identical wave - fronts between the pilot and the traffic signals , hence , allowing the current is - 95 coherent demodulation at the mobile unit . this &# 34 ; beam matching &# 34 ; is facilitated using beam synthesis based on a minimum root mean square approach . this approach allows for +/- 10 degrees phase matching down to - 10 db points , which is sufficient not to degrade the performance of the coherent demodulator at the mobile unit . the coefficients of the individual downlink beams are set as follows : the overhead data ( pilot , synch and paging ) are transmitted through fixed , relatively wide beams . the downlink traffic data beams are set to match the line of bearing as measured by the uplink channel estimator with sufficient width margin to compensate for bearing error ( due to lack of correlation between up and down links ). it should be noted that even relatively wide downlink traffic beams will provide for significant capacity increase . since the angular spread is getting larger as the distance to the base station decreases , the narrow beam width is estimated based on the estimated distance from the bts . this distance is derived from the time delay as measured by a beam director . since the above approach is based on the statistical profile of the scattering region ( various scattering models are considered ), the system must provide for exceptions : at first , the allocated traffic narrow beam is made wider than needed , and as done with the forward power control , it is gradually narrowed based on the frame erasure rate ( analogous to bit error rate ) that is reported on the uplink . in case the frame erasure rate increases , the traffic beam is widened accordingly . this mechanism will also compensate for situations where the uplink angle of arrival ( aoa ) is very different from the downlink aoa . as is evident from the various embodiments illustrated above , the present invention encompasses within its scope many variations . those skilled in the art will appreciate that additional modifications may also be made to the above embodiments without departing from the scope of the invention . accordingly , the true scope of the present invention should not be construed as limited by the details provided above for the purposes of illustration , but should be determined from the following claims and their legal equivalents .