Voltage supply regulated in proportion to sum of positive- and negative-temperature-coefficient offset voltages
The present invention relates to regulated voltage supplies of output voltages which are the sum of (a) a negative-temperature-coefficient voltage proportional to the offset potential across a forward-biased semiconductor junction and (b) a positive-temperature-coefficient voltage proportional to the difference between the offset potentials of two forward-biased semiconductor junctions operated with differing densities of current flow through them. 
     Such supplies are known generally from U.S. Pat. Nos. 3,617,859 (Dobkin etal) and 3,887,863 (Brokaw). These prior art supplies are not amenable to operation for supplying regulating potentials smaller than bandgap potential or to operation with supply voltages of less than 5 volts or so. The present invention permits substantially greater latitude in the design of regulated voltage supplies. 
     A differential-input amplifier configuration that provides zero-valued output signal as long as there is a positive-temperature-coefficient potential difference between the voltages being compared is described by Wheatley in U.S. Pat. No. 3,851,241. 
     A regulated voltage supply embodying the present invention includes a degenerative feedback loop in which an amplified error signal provides either series- or shunt-regulation of the output voltage. A negative-temperature-coefficient voltage that is proportional to the offset voltage across a forward-biased semiconductor junction is compared against a portion of the output voltage to develop the error signal. The comparison between these voltages is made in an unbalanced differential-input amplifier configuration that provides a zero-valued error signal so long as there is a positive-temperature-coefficient potential difference between the voltages being compared. The output voltage is proportional to the sum of the negative- and positive-temperature-coefficient voltages. 
    
    
     In the drawing: 
     FIGS. 1 and 2 are schematic diagrams, each of a series voltage regulator embodying the present invention; 
     FIGS. 3 and 4 are schematic diagrams, each of a shunt voltage regulator embodying the present invention; 
     FIG. 5 is a schematic diagram partially in block form of another voltage regulator embodying the present invention; 
     FIGS. 6, 7, 8, 9 and 10 are schematic diagrams, each showing a pair of transistor means suited for asymmetrical differential-input amplifier operation in accordance with the present invention; and 
     FIGS. 11 and 12 are schematic diagrams of, respectively, a series voltage regulator and a shunt voltage regulator, each embodying the present invention. 
    
    
     Referring to FIG. 1, the series voltage regulator 100 develops a regulated output voltage, V OUT , between its terminals 101 and 102. A potential divider 103 comprising resistors 104 and 105 applies a voltage substantially equal to V OUT  R 105  /(R 104  +R 105 ) to terminal 106. The resistances R 104  and R 105  of resistors 104 and 105, respectively, are assumed to be small compared to the load imposed on the potential divider terminal 106. 
     Current source 107 supplies a substantially constant forward current, I 108 , to semiconductor junction diode 108 (which may be a self-biased transistor). Assuming I 108  to be constant despite changes in temperature, the resulting offset voltage V 108  will (as is well-known) exhibit a negative-temperature-coefficient. A potential divider 109 comprising resistors 110 and 111 applies a voltage V 108  R 111  /(R 110  +R 111 ) to terminal 112. The resistances R 110  and R 111  of resistors 110 and 111, respectively, are assumed to be small compared to the load imposed on terminal 112 and be large as compared to the resistance offered by diode 108. 
     Terminals 106 and 112 are the input terminals of a differential-input amplifier 113 which supplies an output current from terminal 114. This output current assumes a substantially zero value when the difference between the potentials at terminals 106 and 112 has a particular positive-temperature-coefficient value. The emitter-coupled transistors 115 and 116 have their respective base electrodes connected to terminals 112 and 106, respectively. Transistors 115 and 116 have base-emitter junctions with similar profiles and areas in m:l ratio as indicated by the encircled m and l near their respective emitters. Constant current source 117 determines the sum of the respective emitter currents I E115  and I E116  of transistors 115 and 116, responsive to which currents transistors 115 and 116 demand respective collector currents I C115  and I C116 . I C115  is amplified by a factor of -n by the current mirror amplifier 118 comprising transistors 119 and 120. The base current I B121  supplied to common-collector transistor 121 is, then, defined by the following equation. 
     
         I.sub.B121 = (-n) (-I.sub.C115) - I.sub.C116  = nI.sub.C115 - I.sub.C116 (1) 
    
     the values of I C115  and I C116  can be determined as a function of the differences, ΔV, between the base potentials of transistors 115 and 116 by proceeding from the following equation descriptive of transistor action. 
     
         V.sub.BE = (kT/q)ln(K.sub.C /AJ.sub.S)                     (2) 
    
     wherein 
     V be  is the base-emitter potential of the transistor, 
     k is Boltzmann&#39;s constant, 
     T is the absolute temperature at which the transistor is operated, 
     q is the charge on an electron, 
     I c  is the collector current of the transistor, 
     A is the effective area of the base-emitter junction of the transistor, and 
     J s  is the value of I C  /A when V BE  = 0. 
     Transistors made concurrently by the same processing steps in a monolithic structure have substantially equal J S  &#39;s and, if located proximate to each other in the structure, have substantially equal T&#39;s. The quantities V BE , I C  and A will bear the subscripts of the transistors to which they relate in equations 3 and 4, which particularize equation 2. 
     
         V.sub.BE116 = (kT/q)ln(I.sub.C116 /A.sub.116 J.sub.S)      (3) 
    
     
         v.sub.be115 = (kT/q)ln(I.sub.C115 /A.sub.115 J.sub.S)      (4) 
    
     equation 5 is derived by substituting equations 3 and 4 into the defining equation 5 for ΔV. 
     
         Δv = v.sub.be116 - v.sub.be115                       (5) 
    
     
         Δv = (kT/q)ln(I.sub.C116 /I.sub.C115)(A.sub.115 /A.sub.116)(6) 
    
     for I B121  = 0, I C116  = nI C115  ; and A 115  /A 116  = m. Substitution of these values into equation 6 defines the value of ΔV for which I B121  = 0. 
     
         Δv = (kT/q)lnmn                                      (7) 
    
     By choosing the product of m times n larger than unity, the ΔV for maintaining the error signal applied as base current to transistor 121 substantially zero will obtain when V BE116  exceeds V BE115 . 
     The emitter of transistor 121 is connected to terminal 102 completing a degenerative feedback loop comprising potential divider 103, the differential-input amplifier 113 and transistor 121 connected as a common-collector amplifier. The error signal applied as base current I B121  to transistor 121 is not exactly zero-valued but has the small positive value required to support an emitter current I E121  from transistor 121 sufficient to develop the potential V OUT  across the combined resistances of resistors 104 and 105. Supposing the common-emitter forward current gain of transistor 121 to be reasonably large (and, if necessary, transistor 121 can be replaced by a Darlington cascade connection of two or more transistors to make this so) I B121  will be relatively small compared to nI C115  and to I C116 . The degenerative feedback loop will tend to be in equilibrium, then, when the divided-down potential V 106  = R 105  V OUT  /(R 104  +R 105 ) applied to terminal 106 equals the divided down potential V 112  = R 111  V 108  /(R 110  + R 111 ) at terminal 112 plus the (k T/q)ln(mn) potential required between terminals 112 and 106 to maintain nI C115  substantially equal to I C116 . The degenerative feedback loop will, then tend to maintain the following V OUT  between terminals 101 and 102. 
     
         V.sub.OUT = [(R.sub.104 +R.sub.105)/R.sub.105 ] · [R.sub.111 V.sub.108 /(R.sub.110 +R.sub.111)]+(kT/q)ln(mn)           (8) 
    
     Analyzing equation 8 one observes V OUT  is the sum of the negative-temperature-coefficient potential R 111  (R 104  +R 105 ) V 108  /R 105  (R 110  +R 111 ) and the positive-temperature-coefficient potential [(R 104  +R 105 )kT/q R 105  ]ln(mn). By choosing appropriate values of R 110 , R 111 , m and n, V OUT  can be made to be temperature-independent. 
     A V OUT  appreciably larger than that of equation 8 will increase the V 105  = R 105  V OUT  /(R 104  +R 105 ) potential applied to terminal 106. Its increased base potential vis-a-vis that of transistor 115 causes transistor 116 to demand a larger I C116 , while I C115  is reduced. Since I B121  = nI C115  - I C116 , I B121  is reduced. Responsive to the reduced I B121 , I E121  is reduced. The reduced current flow through the resistors 104 and 105, reduces the potential drop due to I E121  developed across them. V OUT  is accordingly reduced towards the value set forth in equation 8. On the other hand, a V OUT  appreciably smaller than that of equation 8 reduces the base potential of transistor 116, reducing I C116  vis-a-vis I C115 . Since I B121  = nI C115  - I C116 , I B121  and consequently I E121  is increased. The increase of I E121  increases the potential drop V OUT  across the resistors 104 and 105. 
     In regulator 100, voltage supply 122, shown as a battery, supplies a positive operating potential B+ to the common connection of current mirror amplifier 118 and the collector electrode of transistor 121. Current source 107, for example, simply may comprise a resistive element, the current flow therethrough being by Ohm&#39;s Law equal to the potential drop thereacross, the B+ potential less the offset potential across diode 108, divided by its resistance. Current source 117 may be provided by a resistor connecting the interconnection of the emitter electrode of transistors 115 and 116 to a point of negative potential or by the collector-to-emitter path of a further transistor biased for constant current operation. 
     FIG. 2 shows a series voltage regulator 200 wherein means, alternative to current mirror amplifier 118, responds to the difference between fixed proportions of I C115  and I C116  to supply an error signal. The alternative means comprises a current mirror amplifier 223 including transistor 224 and 225. I C116  is applied to the input connection of the current mirror amplifier 223, to cause a response current I C116  /n to flow as collector current from transistor 225. Common-emitter amplifier transistor 226, the collector electrode of which is galvanically connected to output terminal 102, thus experiences as an error signal a base current I B226  defined as follows in accordance with Kirchoff&#39;s Law of Currents. 
     
         I.sub.B226 = (I.sub.C116 /n) - I.sub.C115                  (9) 
    
     the inverting nature of its common-emitter amplifier configuration causes transistor 226 to supply a collector current proportional to nI C115  - I C116 , as before. In this connection, the scaling factor n reduces feedback loop gain rather than increasing it as with the FIG. 1 configuration. However, the FIG. 2 configuration has the attractive feature that the action of its feedback loop to maintain the collector potentials of current mirror amplifier transistors 224 and 225 substantially equal, forestalling departures of the current gain of amplifier 223 from 1/n due to &#34;Early effect&#34;. Feedback loop gain in regulators per FIG. 2 can be increased by replacing transistor 226 with a compound device comprising an initial common-emitter amplifier transistor followed in cascade by at least one common-collector amplifier transistor of complementary conductivity type, as illustrated in FIG. 11. 
     FIG. 3 shows a shunt voltage regulator 300 wherein an n( C115  - I C116  error signal is applied to the base electrode of transistor 329. Responsive to the error signal transistor 329 exhibits decreased conduction if the voltage V OUT  appearing between terminals 101 and 102 decreases below the value in equation 8. Decreased conduction of transistor 329 lessens the flow of current through dropping resistor 330, increasing V OUT . The error signal will increase the conduction of transistor 329 if V OUT  increases above the value in equation 8, and the resulting increased flow of current through dropping resistor 330 will increase the voltage drop thereacross, reducing V OUT . Dropping resistor 330 might alternatively be connected between the common connection of current mirror amplifier 118 and terminal 102, rather than the B+, positive connection of supply 122. The connection shown avoids the influence of Early effect on the current gain of current mirror amplifier 118 by arranging transistors 119 and 120 to have substantially like-valued collector potentials. 
     FIG. 4 shows a shunt voltage regulator 400 wherein a (I C116  /n) - I C115  error signal is applied to the base electrode of transistor 431. Responsive to this error signal, transistor 431 exhibits decreased or increased conduction depending on whether the voltage V OUT  between terminals 101 and 102 decreased below or increases above the value in equation 8. Decreased conduction of transistor 431 when V OUT  decreases below the equation 8 value, decreases the current flow through dropping resistor 432; the resulting decreased potential drop across resistor 432 tends to increase V OUT  to the equation 8 value. Increased conduction of transistor 431 when V OUT  increases above the equation 8 value increases the current flow through dropping resistor 432; the resulting decreased potential drop across resistor 432 tends to decrease V OUT  to the equation 8 value. 
     FIG. 5 shows a voltage regulator 500 using still another means for responding to the difference between fixed proportions of I C115  and I C116  to supply an error signal. In accordance with ohm&#39;s Law, I C115  causes a potential drop nRI C115  across resistor 533, which has a resistance R 533  n times as large as the resistance R 534  of resistor 534. In accordance with Ohm&#39;s Law, I C116  causes a potential drop R 534  I C116  across resistor 534. The nR 534  I C115  and R 534  I C116  potential drops are applied to the inverting and non-inverting terminals, respectively, of a differential-input amplifier 535 (e.g., an operational amplifier) which supplies an output current to terminal 102 which is an amplified response proportional to I C116  - nI C115 . 
     Voltage regulator configurations similar to 100, 200, 300 or 400 of FIGS. 1-4, respectively, but in which current mirror amplifiers 118 and 223 are replaced by known equivalent current mirror configurations and in which any necessary modifications of the biasing of the emitter electrodes of transistors 226 and 329 needed to accomodate these changes are also feasible. 
     FIGS. 6, 7, 8, 9 and 10 show configurations 600, 700, 800, 900 and 1000 that each comprise a pair of transistor means that can replace the emitter-coupled pair of transistor 115 and 116 for generating the I C115  and I C116   currents that are differentially combined in predetermined proportion n to develop the error signal in voltage regulators embodying the present invention. Transistors 636 and 637 have their collector electrodes biased to condition each of them for common-collector amplifier operation. Configuration 600 causes a value m in equation 7 equal to the product of p times q times r where: p is the ratio of the effective area of the base-emitter junction of transistor 636 to that of transistor 637; q is the ratio of the current I 638  supplied by source 638 to the emitter electrode of transistor 637, divided by the current I 639  supplied by source 639 to the emitter electrode of transistor 636; and r is the ratio of the base-emitter junction of transistor 640 to that of transistor 641. Elements 637, 638 and 640 comprises transistor means for generating I C115  and elements 636, 639 and 641 comprise transistor means for generating I C116 . 
     Configuration 700 of FIG. 7 causes a value of m in equation 7 equal to the product of p times s where: p is the ratio of the effective area of the base-emitter junction of transistor 636 to that of transistor 637, and s is the ratio of the effective area of the base-emitter junction of transistor 742 to that of transistor 743. To this end, source 744 supplies the combined base current demands of transistors 742 and 743 from a relatively high source impedance compared to that presented by transistors 742 and 743 at the interconnection of their base electrodes. Elements 636 and 742 comprise transistor means for generating I C115 , and elements 637 and 743 comprise transistor means for generating I C116 . Configurations 600 and 700 provide the bases for unbalanced differential-input amplifiers that achieve larger values of m, permitting smaller division ratios for potential divider 109 and thus enhanced accuracy in the determination of larger zero-temperature-coefficient V OUT  potentials, without taking up excessive area in a monolithic integrated circuit. 
     More savings of area on the monolithic circuit can be obtained through the use of further diodes as in configurations 800 and 900 of FIGS. 8 and 9, if the m to be realized be sufficiently large. These additional diodes can be simple PN junctions, or self-biased transistors, but Schottky barrier diodes (e.g., of platinum-silicide-silicon types) can reduce the absolute values of the offset voltages necessary to obtain desired differences in relative offset voltages. Configuration 800, like configuration 700, is advantageous in that the same currents that flow through transistors 742 and 743 flow through transistors 636 and 637, respectively, to conserve current drain on the energizing supply. Configuration 800 causes a value of m in equaton 7 equal to p times s times t u  where: u is the number of diodes in each of serial connections 845 and 846 of diodes, and t is the effective area of the diode junctions in serial connection 845 relative to that of the diode junctions in serial connection 846. Elements 636, 845 and 742 comprise transistor means for generating I C115  ; and elements 637, 846 and 743 comprise transistor means for generating I C116 . 
     Configuration 900 of FIG. 9 causes a value of m in equation 7 equal to p times q (u+1) times t u  times v w  times r where: w is the number of diodes in each of the serial connections 947 and 948, and v is the effective area of the diode junctions in serial connection 947 relative to that of the diode junctions in serial connection 948. It is relatively easy to generate large enough values of m in configuration 900 to permit one to apply the full offset potential developed across diode 108 to terminal 112; as shown, if desired, in effect making R 111  /(R 110  + R 111 ) unity valued in equation 8. In configuration 900, elements 637, 846, 638, 640 and 947 comprise transistor means for generating I C115  ; and elements 636, 845, 639, 641 and 948 comprise transistor means for generating I C116 . Placing elements 637 and 846 in one current path and elements 640 and 947 in another reduces the proportion of the B+ operating potential needed for getting a given number of junctions into the process of determining the offset potential of this transistor means versus the other. 
     Use of still further current paths in each transistor means as shown in FIG. 10 can reduce the proportion of the B+  operating potential needed. There will be an associated tendency towards self-oscillation in the voltage regulator which, as appreciated by workers of average skill in the art, can be controlled by by-passing the feedback loop with a suitably placed capacitance. 
     Configuration 1000 of FIG. 10 causes a value of m in equation 7 equal to the product pqrabcd where: a is the ratio of the effective area of the base-emiter junction of transistor 1049 to that of transistor 1050; b is the ratio of the current I 1051  withdrawn by source 1051 from the emitter electrode of transistor 1050, divided by the current I 1052  withdrawn by source 1052 from the emitter electrode of transistor 1049; c is the ratio of the effective area of the base-emitter junction of transistor 1053 to that of transistor 1054; and d is the ratio of the current I 1055  supplied by source 1055 to the emitter electrode of transistor 1054, divided by the current I 1056  supplied by source 1056 to the emitter electrode of transistor 1053. Only about a volt of operating potential need be apportioned to this configuration. Elements 637, 638, 1049, 1052, 1054, 1055 and 640 comprise transistor means for generating I C115  ; and elements 636, 639, 1050, 1051, 1053, 1056 and 641 comprise transistor means for generating I C116 . 
     FIG. 11 shows a series voltage regulator 1100 illustrating how one can handle various incidental problems concerning the design of voltage regulators embodying the present invention. The negative-temperature-coefficient potential V 108  is developed at the base electrode of grounded-emitter amplifier 1157 to adjust its collector current I C1157  substantially to equal in amplitude a temperature-independent current I C1158  supplied from the collector electrode of transistor 1158. The error signal current by which I C1158  exceeds I C1157  in amplitude is applied to the base electrode of emitter-follower transistor 1159, used to provide the direct-coupled collector-to-base feedback connection for transistor 1157. In this arrangement, emitter-follower transistor 1159 acts as a buffer amplifier, greatly reducing the tendency towards I C1158  being diverted in part to the resistive potential divider 109, as would happen if the collector-to-base feedback connection of transistor 1157 were provided by direct connection rather than by emitter-follower transistor 1159. 
     I C1158  is temperature-independent if a proper value temperature-independent voltage is applied across the series connection of its base-emitter junction and its positive-temperature-coefficient emitter degeneration resistor 1160. This voltage is obtained by scaling from the temperature-independent potential V OUT  in a resistance amplifier essentially consisting of transistor 1161, with emitter-resistance provided by the series connection of resistors 104 and 105 of potential divider 103, and with collector resistance provided by resistor 1162 across which appears the temperature-independent potential sought for application to elements 1158 and 1160. This potential is also applied across the series connection of the base-emitter junction of transistor 1163 and its emitter degeneration resistor 1164, to generate the I 638  current flow from its collector electrode, and across the series connection of the base-emitter junction of transistor 1165 and its emitter degeneration resistor 1166, to generate the I 639  current flow from its collector electrode. 
     Transistor 1161 is in cascade connection with transistor 226 to improve the loop gain of the basic voltage regulating function determining V OUT . V 108  is applied to the base electrode of grounded-emitter transistor 1167 to bias it for placing constant collector current demand on the joined emitter electrodes of transistors 640 and 641. Resistor 1168 shunting the collector-to-emitter path of transistor 1161 is used to prevent an undesired equilibrium condition in the regulator 1100 wherein none of the transistors is conductive, necessary where, as here, current levels in the circuit are determined regeneratively. 
     FIG. 12 shows a shunt voltage regulator 1200, which type of regulator is attractive for monolithic integrated circuits where the dropping resistor 330 as well as supply 122 are located off-circuit. Attractive, because the power dissipation in the resistor 330 does not heat up the integrated circuit. In regulator 1200 resistor 104 is in two parts 104a and 104b, the potential drop across resistor 104a being used to bias transistor 1158 for supplying a constant collector current, I C1158 . I C1158  is supplied to the collector of transistor 1157 primarily via the collector-to-emitter 104a  path of transistor 1267. Transistor 1267 is conditioned to pass this current by direct-coupled collector-to-base feedback, the connection being made via emitter-follower transistor 1268 and the forward-biased emitter-base junction of transistor 1269. The base current I B1269  of transistor 1269 is equal in amplitude to the base current I B1267  of transistor 1267 required for regulating the collector current I C1267  of transistor 1267 so as to substantially equal I C1158  in amplitude. The collector current I C1269  of transistor 1269 is applied to the emitter electrode of similar type transistor 1270 to cause a base current I B1270  to flow from transistor 1270, which I B1270  is substantially equal to I B1269  and thus to I B1267 . The current is divided between the base electrodes of transistors 742 and 743 to proportion their collector currents to I C1158 . 
     A wide variety of voltage regulators may be constructed based on the principles of the present invention, as should be apparent from the numerous examples set forth above, and the scope of the following claims should be accordingly construed.