Improved pulse-width-modulation signal converter
BACKGROUND OF THE INVENTION 
     This invention relates to the field of electronic measurement and more particularly to a measuring circuit utilizing operational amplifiers to convert an analogue signal into a pulse-width modulated signal which can then be transmitted without information loss to a remote unit for measurement. The signal converter has a first integrator whose summing point is connected to the signal input and also connected to the output of a switching element; and which is then connected in series to a threshold value controller whose output signal actuates the switch element, which in turn provides a positive or negative reference voltage to the summing point and the threshold value controller. 
     A type of signal converter is disclosed in German published patent application No. 2,135,802. Such signal converters are required for the further processing of potential-separated measured values, e.g., measured values of armature current or of rpm set and actual values in drive designs, when there can be no losses in the information content. In that arrangement the signal is converted into a frequency proportional to the measured value and then transmitted without any voltage information change. On the low voltage side, this frequency is again converted into a voltage proportional to the frequency. It has been shown that particularly in cases of high sampling rates the linearity of this known signal converter is inadequate so that an offset equalization is required, at least in order to avoid expensive, rapid-acting, precision amplifiers to realize the integrator. 
     In is an object of this invention to provide a signal converter of the type specified above possessing adequate linearity utilizing less expensive operation amplifiers. 
     SUMMARY OF THE INVENTION 
     Briefly stated in accordance with one aspect of the invention, the foregoing object is achieved by providing a signal converter having a first integrator with a summing point connected to: the signal to be measured as an input, the output of a reversing switch element, and in series with a threshold value controller whose output signal activates a reversing element. The reversing element, depending upon the threshold controller, supplies the first integrator summing point and the threshold controller with a positive or a negative reference voltage. Also connected to the signal input is the summing point of a second integrator. The second integrator summing point is also connected to the output of the reversing switch. The output of this second integrator forms a reference signal for the first integrator. 
     Since in the design according to this invention a correcting signal is generated by the second integrator which is supplied to the first integrator until it has been assured that the output signal corresponds precisely to the input signal, the linearity of said signal converter is determined exclusively by the parameters of the second integrator. 
     Preferrably, the output of the first integrator is connected to a third integrator whose output signal is superimposed on the reference signal for the threshold value controller. By use of this correcting voltage, the voltage time waveforms in one cycle of the triangular or sawtooth voltage of the first integrator are corrected to the same magnitude so that there is no further necessity for an offset equalization. The symmetry of the triangular voltage of the first integrator is thus exclusively dependent upon the parameters of the third integrator. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     While the specification concludes with claims particularly pointing out and distinctly claiming the subject matter which is regarded as the invention, it is believed that the invention will be better understood from the following description of the preferred embodiment taken in conjunction with the accompanying drawings: 
     FIG. 1 is a simplified schematic drawing of the signal converter circuit; 
     FIG. 2 is a dual plot of integrator voltage, V 3 , and output voltage V 9  as a function of time for /+V ref  /=/-V ref  / and V 1  =0. 
     FIG. 3 is a dual plot of integrator voltage, V 3 , and output voltage, V 9 , as a function of time for /+V ref  /=/-V ref  / and V 1  &gt;0. 
     FIG. 4 is a dual plot of integrator voltage, V 3 , and output voltage, V 9 , as a function of time for /+V ref  /=/-V ref  / and V 1  &lt;0. 
    
    
     DESCRIPTION OF A PREFERRED EMBODIMENT 
     The signal V 1  is supplied at the input terminals 1 and passes through a resistance 2 to the summing point 3a of a first integrator 3, which in the model is designed with an operation amplifier 3b wired to a capacitor 3c. The output 3d of integrator 3 is connected to the non-inverting input of a threshold controller 4. Output 4a is connected to the actuating inputs of two reversing switch elements 5 and 6 supplied at their alternating contacts 5a and 5b or 6a and 6b by reference voltages +V ref  and -V ref . preferrably of the same magnitude, but with different polarity signs. The output of reversing switch 6 is connected over a resistance 7 to the inverting input of comparator 4, and the output of the reversing switch element 5 is connected over a resistance 8 to the summing point 3a of integrator 3. The output of the reversing switch 5 is, moreover, connected to the terminals 9 at which the pulse width modulated output signal V 9  is obtained. 
     The output 10a of a second integrator 10 is connected to the non-inverting input of integrator 3. Integrator 10 is similarly designed with an operational amplifier 10b, which in normal fashion is wired to a capacitor 10c. The summing point 10d is connected, as in the case of integrator 3, by a resistance 11 to the input terminals 1 and by a resistance 12 to the output of the reversing switch element 5. 
     The output of integrator 3 is connected in series to the summing point 14 a of a third integrator 14 by a resistance 13. Integrator 14 is similarly designed with an operational amplifier 14b which is wired to a capacitor 14c and whose non-inverting input is connected to the reference potential. The output 14d of the integrator 14 is connected by a resistance 15 to the inverting input of the threshold controller 4. 
     The operating principle of the signal converter is further explained by means of FIGS. 2 through 4. These figures plot respectively the output voltage V 3  of integrator 3 and the pulse width modulated output voltage V 9  of the converter over time period 5, wherein the preferable design is assumed to be /+V ref  /=/-V ref  /. It is further assumed that for FIG. 2 V 1ref  =0, for FIG. 3, V 1  =0+ΔV 1 , and for FIG. 4, V 1  =0-ΔV 1 . At time point zero the reversing switch elements 5 and 6 are in the position shown in FIG. 1. Thereby, the voltage +V ref  is supplied to the summing point 3a of integrator 3 and reference voltage -V ref  supplied to the inverting input of comparator 4. Integrator 3 integrates the voltage prevailing at summing point 3a until its output voltage V 3  attains the value +V ref . At that point the threshold value controller 4 flips over and reverses the reversing switch elements 5 and 6. Integrator 3 now integrates a negative voltage prevailing at summing point 3a until its output voltage V 3  attains the value -V ref  at which the comparator 4 again flips over and reversing switch elements 5 and 6 return to the position shown in FIG. 1, and the cycle resumes. Thereby, one obtains the triangular voltage V 3  shown in FIG. 2 and the output voltage V 9 . With the precondition that /+V ref  /=/-V ref  /, the time span t 1  in which the integrator 3 integrates downward, equals the time space t 2  in which it integrates upwards so that t 1  /t 2  =1. The output voltage alternating between +V ref  and -V ref  thus has the arithmetic average value of zero (V 9  =0), which corresponds to the input signal V 1  =0. 
     In FIGS. 3 and 4, V 1  &gt;0 or V 1  &lt;0. Thus, the time span t 1  for the downward integration is different from the time span t 2  for the upward integration and t 1  /t 2  ≠1. Thereby, t 1  &lt;t 2  for V 1  &gt;0 and t 1  &gt;t 2  for V 1  &lt;0. The arithmetic average of the output voltage V 9  is thus proportional to the input voltage. As the duty cycle of the output voltage V 9  is evaluated solely, the clock frequency and thus also the capacitance of capacitor 3c has no effect on the precision of this conversion process. 
     Particularly in case of high clock frequencies component tolerances can produce transmission errors. In order to compensate for these transmission errors, integrator 10 generates a correcting signal which serves as a reference voltage for integrator 3 and follows it until V 9  =V 1 . Preferably, the time constant selected for that process determined by resistance 11 and capacitor 10c is large relative to the clock frequency. The linearity between the input voltage V 1  and the output voltage V 9  is thus dependent exclusively on the parameters of amplifier 10b, particularly in those cases when V 1  is symmetrical to the reference point. 
     Integrator 14 generates a correcting voltage which is superimposed on the reference voltage for comparator 4 and thereby shifts the operational amplifier 4 that the voltage-time waveforms F 1  and F 2  in FIGS. 2 through 4 always remain of the same magnitude. The symmetry of V 3  is thus exclusively dependent on the parameters of the operation amplifier 4. 
     These corrections have a particular impact in case of high sampling rates, e.g., at clock frequencies of 100 kHz required for measurements with high resolution, of 1:10.000 to 1:40.000. Under these requirements it would be necessary with known signal converters to install very rapid-acting, precision amplifiers as the operational amplifiers 3b and 4, which would, no doubt, be very expensive. Conversely, in the circuitry as described the operational amplifiers 10b and 14b can be utilized as slow-acting precision amplifiers, while the operational amplifiers of integrator 3 and threshold value controller 4 could be rapid-acting operational amplifiers whose precision does not have to be excessively high. At a slight expense, therefore, with the circuitry in accordance with this invention achieves a slight offset (e.g., less than 100 micro volts) which requires no equalization while attaining high linearity (e.g., greater than 14 BIT) with high information sampling rates. 
     As will be evident from the foregoing description, certain aspects of the invention are not limited to the particular details of the examples illustrated, and it is therefore contemplated that other modifications or applications will occur to those skilled in the art. It is accordingly intended that the claims shall cover all such modifications and applications as do not depart from the true spirit and script of the invention.