Differential reference voltage generator for NMOS single-supply integrated circuits
FIELD OF OF THE INVENTION 
     The present invention relates to integrated circuit technology and more particularly it concerns a differential reference voltage generator for NMOS single-supply integrated circuits. 
     BACKGROUND OF THE INVENTION 
     In integrated analog circuits in NMOS technology (n-channel MOS) with a single voltage supply having not high level (e.g. 5 V), some circuit parts have limited output voltage swing and require a differential reference voltage, i.e. two reference voltages for minimum and maximum signal levels, since minimum signal level is different from ground; besides the difference between the two reference voltages is to remain stable. 
     Examples of such circuit parts are analog-to-digital or digital-to-analog converters where the weighting network output is decoupled by a voltage follower amplifier whose output voltage swing is limited and requires a voltage reference different from ground for minimum signal level. 
     NMOS single-supply circuits for generating single reference voltages are already known in the art, as that described in the paper &#34;A new NMOS Temperature-Stable Voltage Reference&#34; by R. A. Blauschild et al., IEEE Journal of Solid-State Circuits, vol. SC-13, pp. 767-774, December 1978. 
     In said circuit the reference voltage is derived from the difference between gate-source threshold voltages of an enhancement and a depletion MOS transistors both implemented with the same technology. Reference voltage is kept stable by a feedback obtained with a high-gain differential amplifies; that gives rise to serious stability problems, making it necessary to insert a compensating network, for the feedback loop, taking up a large silicon area. Moreover, reference voltage has a fixed and not-programmable value. 
     OBJECTS OF THE INVENTION 
     These problems are overcome by the present invention of a differential reference voltage generator which does not require a high-gain feedback loop to compensate for thermal drift, and wherein the differential voltage is maintained stable by annulling the difference between temperature-dependent terms in the equations of voltages and currents of the network which generates said differential voltage. 
     Besides the mean value of differential voltage can be varied with respect to ground. 
     It is a particular object of the present invention the device described in claim 1. 
    
    
     SPECIFIC DESCRIPTION 
     The characteristics of the present invention will be now described with reference to a non-limiting example thereof, in connection with the annexed drawing, wherein the electric diagram of the generator is shown. 
    
    
     In the FIGURE MD1, MD2 denote two MOS depletion transistors, whose drains are connected to supply voltage V DD , and whose gates are connected to one another and to MD1 source. 
     ME1, ME2 denote two MOS enhancement transitors, whose drains are connected to their respective gates and to the sources of MD1 and MD2 respectively. 
     ME3, ME4 denote two MOS enhancement transistors, which have drains connected to ME1, ME2 sources respectively, gates interconnected, and sources connected to ground. 
     Besides drain and gate of ME3 are interconnected. 
     ME3 and ME4 are connected in &#34;current mirror&#34; configuration, that is why their drain currents have equal value. In addition transistor MD2 is connected in common-drain configuration. 
     Voltage V H  present at the source of MD2 is the higher-level reference voltage. Voltage V L  present at the gate of ME3 is the lower-level reference voltage. 
     Value V DIF  =V H  -V L  is the required differential reference voltage. 
     In the FIGURE all the transistors are n-channel transistors. The general equation which expresses drain current I D  versus gate-source voltage V GS  in a MOS transistor in strong inversion is as follows: 
     
         I.sub.D =β·K(V.sub.GS -V.sub.T).sup.2 /2     (1) 
    
     where β=μ·C ox , which μ[m 2  ·s/V] charge-carrier mobility, and C ox  [F/m 2  ] specific gate capacity; K=W/L with W and L channel cross-section and length respectively; V T  gate-source threshold voltage. 
     The values of current and voltage in the circuit shown in the FIGURE can be calculated by means of equation (1). More particularly current I R1 , which is the drain current of transistors MD1, ME1 and ME3 is: 
     
         I.sub.R1 =β.sub.MD1 ·K.sub.MD1 ·(V.sub.MD1).sup.2 /2 (2) 
    
     where the parameters are those of transistor MD1, whose V GS  is equal to zero, as it results also from the FIGURE. 
     Voltage V L  is gate-source voltage V GSME3  of transistor ME3; making use of equations (1) and (2) we derive: ##EQU1## 
     Voltage V H  will be on the contrary: ##EQU2## 
     There considering that I R1  =I R2 , and that I R2  is the drain current of MD2, ME2, ME4 the result will be: ##EQU3## 
     Differential reference voltage V DIF  depends therefore on the difference of threshold voltages of transistors ME1 and MD2, and on a term which can be kept equal to 0 by dimensioning said transistors so that: 
     
         β.sub.ME1 ·K.sub.ME1 =β.sub.MD2 ·K.sub.MD2 
    
     Therefore by duly dimensioning the transistors whereon value V DIF  depends, terms varying with temperature in equation (5) annul each other. 
     Hence V DIF  is very stable. 
     Voltages V H  or V L  can be set by duly dimensioning transistors ME2, ME3, ME4, so as to exploit as well as possible the output voltage swing of the amplifier which requires the reference voltage generator.