Multiple level voltage comparator circuit
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to circuits used for comparing voltages, and in particular to a voltage comparator circuit for use in the pin electronics circuits of automatic test equipment. 
     2. Description of the Prior Art 
     In automatic test equipment for the testing of integrated circuits, a plurality of pin electronics interface circuits are coupled to the pins or other nodes of an electronic device being tested. Through these pins, stimuli signals are supplied to the device under test, and output signals from the device under test are detected and measured. Usually, the stimuli signals represent logic states or analog voltages or currents which are desired to be impressed on the pins of the device under test as a parallel pattern, with the resulting output signals checked in parallel. 
     The interface circuits function as an interface between the computer controlling the test system and the individual pins of the device under test. The interface circuits receive reference voltages and digital data from other circuits in the test system, and then through a driving switch these voltages or data onto the desired pin of the device under test under control of a program stored in the test system computer. Correspondingly, the interface circuits receive reference voltages or data from the device under test and supply it to a comparator circuit which compares the signal received with the proper response stored in the program of the test system computer. In such apparatus voltage comparators are frequently used to determine the potential of one or more pins of the device being tested relative to one or more reference potentials. 
     As integrated circuits have increased in speed, size, and complexity, the demands made on the voltage comparators in automatic test equipment have correspondingly increased. With the advent of commercially available automatic test equipment operating from a 50 mega-Hertz clock signal, improved voltage comparators are necessary to enable testing of the device under test at these high clock rates. 
     In the prior art, high voltage comparators relied upon balanced differential pairs of field effect transistors for detecting the relative magnitude of a reference voltage with an unknown voltage. Unfortunately, such devices are not suitable for detecting failures of short duration such as those found when testing emitter coupled logic devices. 
     In other prior art voltage comparison circuits, fast low voltage comparators were feasible, but could not handle the voltage swings of TTL and MOS logic families. Typical prior art voltage comparators are shown in: &#34;Sentry® High Speed and Clock Driver Pin Electronics Reference Manual&#34; available from Fairchild Test Systems Group, San Jose, California. 
     SUMMARY OF THE INVENTION 
     This invention provides a voltage comparator which, in the preferred embodiment, accepts a signal between minus four volts and plus seven volts and compares that signal to four independent reference potentials. Because the input devices remain in class A at all times, the circuit is fast, and the base-emitter breakdown of the input devices does not limit the differential input voltage. The circuit combines very high speed transistors with a wide voltage range capability. Because the input devices are always on, stored charge remains nearly constant and the change in propagation delay which accompanies overdriving is significantly reduced. 
     In a preferred embodiment a voltage comparator for comparing an unknown voltage relative to a reference voltage comprises first and second input transistors each having a first electrode for receiving a potential, the first electrode of the first transistor connected to a reference voltage and the first electrode of the second transistor connected to the unknown voltage, a second electrode of the first input transistor connected to a first current source, a second electrode of the second input transistor connected to a second current source; a third current source; first and second diodes each having a first electrode coupled to the third current source, the first diode having a second electrode connected to the second electrode of the first transistor, and a second diode having a second electrode coupled to the second electrode of the second transistor; and output detection means coupled to a third electrode of each of the first and second transistors for detecting which is being supplied current from the third current source. 
     In the preferred embodiment the voltage comparator circuit includes two independent voltage comparators, two dependent voltage comparators, and two temperature tracking networks to stabilize the operation of all comparators. To increase operating speed the diodes are preferably Schottky barrier diodes. In one embodiment a serially connected resistor and capacitor are connected between the second electrodes of the first and second transistors. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram of two independent voltage comparators for comparing the unknown voltage with a high and a low reference voltage. 
     FIG. 2 is a schematic of two voltage comparator circuits coupled to the circuit shown in FIG. 1 for comparing the unknown voltage with two intermediate voltages. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIGS. 1 and 2 are schematic drawings of the preferred embodiment of this invention. The circuits shown in FIGS. 1 and 2 are intended to be connected to each other at nodes A, B, C, and D as shown at the right-hand edge of FIG. 1 and the left--hand edge of FIG. 2. 
     The comparator circuits shown in FIGS. 1 and 2 accept an input signal CB01 and compare it to four different reference potentials - comparator reference high &#34;CRH,&#34; comparator reference low &#34;CRL,&#34; comparator reference intermediate high &#34;CRIH,&#34; and comparator reference intermediate low &#34;CRIL.&#34; Two of those potentials, CRH and CRL, have independent comparators associated with them designated first and second comparators in FIG. 1. The comparators associated with the CRIH and CRIL potential levels shown in FIG. 2 are coupled together to provide a single output signal indicative of whether the unknown potential is between the CRIH and CRIL potentials. 
     The inputs to the four comparators are divided to minimize loading on the buffer circuit typically used to supply the signal CB01 to the comparators. There are three differential output pairs, CHO and CHO, CLO and CLO, CIO and CIO. These outputs are intended to drive hundred ohm loads, and be buffered by subsequent gain stages. 
     The structure of each of the four comparators shown in FIGS. 1 and 2 is similar. In the first comparator, transistors Q105, Q106 and Q107 have their bases connected to node C and emitters coupled through resistors R107, R109, and R111, respectively, to node D. The collector of each transistor is coupled through a corresponding resistor R106, R108 and R110. A differential pair of transistors consisting of transistors Q103 and Q104 have their emitters coupled together through resistor R105 and capacitor C103. Diode D101 is connected between the emitter of transistor Q103 and resistor R108, while diode D102 is connected between the emitter of transistor Q104 and resistor R108. A resistor R112 connects the comparator reference high potential CRH to the base of transistor Q103, while a similar resistor R113 connects the base of transistor Q104 and the source of signal CB01. 
     The collector outputs of the differential pair Q103 and Q104 are connected through resistances R101 and R102, respectively, to node A. A differential output pair consisting of transistors Q101 and Q102 have their bases shorted together. The emitter of transistor Q101 is connected to the collector of transistor Q103, while the emitter of transistor Q102 is connected to the collector of transistor Q104. The collector of Q101 supplies output signal CHO, while the collector of transistor Q102 supplies output signal CHO. A compensation network consisting of resistor R103 and capacitor C101 is connected between ground and the emitter of transistor Q101, while a similar network consisting of resistor R104 and capacitor C102 is connected between ground and the emitter of transistor Q102. 
     The structure of the second comparator parallels that of the first comparator. The corresponding active and passive devices in the second comparator are numbered in the same manner as those in the first comparator except that the first digit of each reference numeral is &#34;2&#34; rather than &#34;1.&#34; The bases of the differential input pair of the second comparator are connected between CB01 and the comparator reference low potential CRL. 
     A temperature tracking network includes an operational amplifier having input terminals connected to potential V2 and through resistor R312 to node A, and an output terminal connected to node B. A transistor Q308 has its emitter coupled to an input of amplifier U1, with collector and base shorted together to node B. In a similar manner a second temperature tracking network U2 has an operational amplifier having input terminals connected to potential V3 and through resistor R313 to node D, and an output terminal connected to node C. The corresponding transistor Q309 has its emitter coupled to node D through resistor R313, and its base and collector shorted to node C. 
     FIG. 2 depicts two further comparators, each constructed in the same manner as the first and second comparators illustrated in FIG. 1. The third comparator, shown generally on the left-hand side of FIG. 2, has components numbered to correspond with similar components in the first comparator, except that the first digit of each reference numeral for the parts of the third comparator is a &#34;3&#34;, rather than a &#34;1&#34;. The fourth comparator is also numbered with reference numerals corresponding to the first comparator except that the first digit of each reference numeral of the fourth comparator is a &#34;4&#34;, rather than a &#34;1&#34;. The base of transistor Q303 is coupled to an intermediate high comparator reference potential CRIH as shown, while the bases of transistors Q304 and Q404 are connected to CB01. Finally, the base of transistor Q403 is coupled to the comparator reference intermediate low potential CRIL. The collector electrodes of transistors Q303 and Q404 are connected together, as are the collectors of transistor Q304 and Q403. The common collectors of the pairs drive the differential output stage which includes transistors Q301 and Q302. Transistor Q301 has an emitter connected to the collectors of transistors Q303 and Q404, while the emitter of transistor Q302 is connected to the collectors of transistors Q304 and Q403. Transistors Q301 and Q302 supply output signals CIO and CIO as shown. 
     The operation of all four comparators is similar. With respect to the first comparator, transistors Q105, Q106, and Q107 function as current sources for the input differential transistor pair, Q103 and Q104. If the input potential CB01 is less than CRH, diode D101 will be forward biased and diode D102 reverse biased. The emitter current of transistor Q103 will therefore be the sum of the collector currents of transistors Q105 and Q106, while the emitter current of transistor Q104 will be the collector current of only transistor Q107. Because the voltage across resistors R101 and R102 will be substantially constant, any change in the collector current of transistor Q103 will result in an equal change in collector current for transistor Q101. Similarly a change in collector current for transistor Q104 results in an equal change for the collector current of transistor Q102. Because the emitter resistance of transistor Q101 is very small and is in parallel with resistor R101 almost all of the current change resulting from the input stage goes directly into changing the emitter current of the output stage. 
     Of particular advantage to this circuit is the fact that all transistors are always operating in the high f t  range of their active region. The only devices which actually switch on and off are Schottky diodes D101 and D102, and these devices have recovery times on the order of less than 50 picoseconds. 
     Because the input transistors Q103 and Q104 operate in cascode with the output transistors Q101 and Q102, there is no feedback from output to input. Very high gains are also possible because the Miller capacitance of the input devices are not multiplied by the gain of the output stage. The circuit is also advantageous because its voltage range is limited only by the collector-emitter breakdown voltage of the input devices, typically on the order of 15 volts. Furthermore, because the input devices, transistors Q103 and Q104, are always on, their charge storage is nearly constant. The only change in charge storage results from the switching of the Schottky diodes D101 and D102, and the charge storage of these devices is relatively small, typically one-fifth that of the input transistors. 
     Resistors R103 and R104, and capacitors C101 and C102 provide a compensation network which avoids oscillation of the output stage at high frequencies. Capacitor C104 holds the base potential of transistor Q102 constant at high frequencies. The emitter potential of transistor Q102 is held constant at low frequencies by the output of operational amplifier U1. Capacitors C206 and operational amplifier U2 provide the same functions for transistors Q105, Q106, and Q107. Resistor R105 and capacitor C103 permit the use of slower transistors Q105, Q106, and Q107, and may be omitted if higher quality devices are employed. 
     Transistor Q308, operational amplifier U1 and resistor R312 form a temperature tracking network which maintains constant voltage across the emitter resistors of the output stages of both the first and second comparators. Transistor Q309, resistor R313, and operational amplifier U2 provide the same function for the emitter resistors R107, R109, and R111 of the current source transistors Q105, Q106 and Q107. 
     The circuits shown in FIG. 2 operate in the same manner as that described in conjunction with FIG. 1 except that the outputs of the comparators shown in FIG. 2 are combined to provide a single output signal CIO and its complement CIO. In the preferred embodiment the resistances of R307, R309, and R311 (and R407, R409, and R411) are double the resistances of the corresponding resistors in FIG. 1 to thereby halve the current flow. 
     The two stages of comparators shown in FIG. 2 are summed together such that the collector currents of transistors Q303 and Q404 are maximized and the collector currents of transistors Q304 and Q403 are minimized, when the unknown potential CB01 is between potential CRIL and potential CRIH. Furthermore, resistances R301 and R302 are unequal so that the outputs CIO and CIO are equal when potential CRIH is greater than potential CB01 and when potential CB01 is greater than potential CRIL. 
     In the preferred embodiment the circuit shown in FIGS. 1 and 2 is fabricated using the component values and types shown below. 
     
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Preferred Component Values and Types                                      
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R101    234 ohms       C101      10 pf                                    
R102     234           C102      10                                       
R103     56            C103       5                                       
R104     56            C104      .01 uf                                   
R105     47            C105      .01                                      
R106     56            C206      .01                                      
R107     587           Q101      PNP                                      
R108     56            Q102      PNP                                      
R109     587           Q103      NPN                                      
R110     56            Q104      NPN                                      
R111     587*          Q105      NPN                                      
R112     50            Q106      NPN                                      
R113     50            Q107      NPN                                      
R301     218           U1        741                                      
R302     267*          U2        741                                      
R307    1174           V1        14.5 v                                   
R309    1174           V2        10.0 v                                   
R311    1174           V3        -7.5 v                                   
R312    1174           V4        -12.0 v                                  
R313    1174                                                              
R407    1174                                                              
R409    1174                                                              
R411    1174                                                              
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 *trimmable                                                               
 
    
    
     Using these component values and types the circuit shown in FIGS. 1 and 2 provides an input voltage range which allows CB01 to fluctuate between minus 4 volts and plus 7 volts. The comparator allows high speed operation over a wider voltage range than prior art comparators. 
     The foregoing description of the preferred embodiment of the invention is intended to illustrate the invention rather than limit it. The scope of the invention may be determined from the appended claims.