Over-current prevention circuitry for transistor amplifiers
The present invention concerns protection circuitry for preventing overly large current in the output transistors of an amplifier, in which the output transistors are arranged for Class B or AB operation, thereby to avoid thermal over-dissipation and secondary breakdown of the output transistors. 
     Burwen and Sondermeyer describe in U.S. Pat. Nos. 3,500,218 and 3,536,958, respectively, the limiting of the permissible base current drives to each of the output transistors between upper and lower values in linear relationship to the decrease and increase of its emitter-to-collector potential between saturation and quiescent value, the lower value of limiting being maintained as the transistor emitter-to-collector potential increases beyond its quiescent value. This is done to prevent operation of the output transistors outside their Safe Operating Area Rating (SOAR). The SOAR is the region of the transistor collector characteristic over which it can be operated without risk of damage or distraction due to over-dissipation or secondary breakdown. 
     The Burwen and Sondermeyer circuits are circuits in which, like earlier circuits, the output transistor to be protected is provided with an emitter resistor for sensing the magnitude of its emitter current flow, and a clamping transistor, the emitter-to-collector path of which is connected across the base-to-emitter path of the output transistor. The voltage developed across the resistor biases the clamping transistor into conduction when the output transistor attempts to draw excessive emitter current. This diverts base current flow from the output transistor. 
     An undesirable feature of these over-current prevention circuits using collector- or emitter-current sensing resistors in series with the collector-to-emitter paths of the output transistors is the following. The potential drops which must appear across the sensing resistors at higher current levels in order to activate the clamping transistors restricts the output signal swing across the load in some degree, reducing available output power. This reduction in output power delivering capability is particularly significant if the direct operating potential available to the amplifier is restricted. Also, the lost power must be dissipated from the amplifier circuitry, which is undesirable in integrated constructions since it tends towards raising the operating temperature of the amplifier. 
     Each of the over-current prevention circuits described above includes a non-linear current feedback circuit for diverting base current from the output transistor when the latter attempts to operate outside its SOAR. 
     Another approach which can be taken is to limit the maximum base current available to the output transistor at any time without recourse to current feedback and then to modulate the flow of this current to the output transistor as a function of input signal. This alternative approach avoids the problems associated with current-sensing resistors. A similar approach is described by Leidich in U.S. Pat. No. 3,855,540, where there is a fixed constraint upon the combined base currents available to a pair of Class AB amplifier output transistors driven in push-pull. The prior art approaches that restrict the availability of drive currents to the output transistors, but are unresponsive to the signal levels in the output transistors, do not permit Class B operation or Class AB operation approaching Class B operation, particularly at power levels of more than a few hundred milliwatts. 
     The present inventor has found that Class B operation or Class AB close to Class B operation can be achieved despite restriction of base drive to the output transistors independent of their collector-to-emitter currents, by making these restrictions depend upon the collector-to-emitter potentials of the transistors. One can achieve operation of the output transistors over portions of their SOAR&#39;s comparable to the portions over which output transistors in the Sondermeyer and Burwen circuits operate while avoiding the need for current sensing resistors. 
    
    
     In the drawing: 
     each of FIGS. 1 and 3 is a schematic diagram of a Class B amplifier embodying the present invention and having output transistors afforded over-current protection as outlined in the previous paragraph; and 
     FIG. 2 is the collector characteristic of an amplifier output transistor having superimposed on it the load-line limiting characteristic provided by the present invention. 
    
    
     In FIG. 1, transistors 30 and 31 are in Darlington-like compound transistor structure, operated as a common-emitter amplifier to amplify low-level signal currents applied to terminal 35 and provided with a collector load comprising, in addition to a multiple-V BE  potential offset regulator comprising elements 32-34, a constant current source at the collector electrode of 36. This constant current source and the relatively small combined quiescent base currents of transistors 11 and 13 supply the quiescent collector current demands transistors 30 and 31 present when an overall degenerative feedback connection around the amplifier (not shown), usually including a preamplifier stage, is completed from terminal 21 to terminal 35. The collector current variations of transistors 30 and 31 are coupled through elements 32-34 to be applied to the joined base electrodes of transistors 11 and 13. 
     Transistors 11 and 12 are connected in an emitter-coupled differential amplifier configuration, as are transistors 13 and 14. The joined emitter electrodes of 11 and 12 are supplied a bias current I 17  from a controlled current source 17, and the joined emitter electrodes of 13 and 14 are supplied a bias current I 19  from a controlled current source 19. Controlled current source 17 limits the collector current transistor 11 can deliver to one of the amplifier output transistors, a compound transistor structure 18. Controlled current source 19 limits the collector current transistor 14 can deliver to the other of the amplifier output transistors, a compound transistor structure 20. Compound transistor structures 18 and 20 are connected to terminal 21 for supplying an external load means, not shown, and the serial connection of their output circuits between operating potential terminals 27 and 28 causes them to provide active internal load means for each other absent connection to an external load means. 
     The base electrodes of transistors 12 and 14 are offset by a potential that is proportional to the ΔV BE  difference in the base-emitter potentials of two transistors operated at substantially the same temperature as transistors 11, 12, 13 and 14. Under quiescent conditions, the potential at the joined base electrodes of 11 and 13 is intermediate to the potentials at the base electrodes of 12 and 14 owing to the overall feedback connection around the amplifier attempting to adjust compound transistor structures 18 and 20 for substantially equal quiescent conduction. Thus, transistor 11 is biased to be substantially less conductive than 12, and transistor 14 is biased to be substantially less conductive than 13. Accordingly, transistors 11 and 14 supply small input currents to the similar compound transistor structures 18 and 20, respectively, to bias them to conduct an idling current that biases them for operating Class AB and thus avoids cross-over distortion. 
     A substantial positive signal current variation applied to the joined base electrodes of transistors 11 and 13 cuts off transistor 11 and reduces conduction of 13. The non-conduction of 11 interrupts the flow of input current to compound transistor structure 18, rendering it non-conductive. The reduced conduction of 13 permits increase in the portion of I 19  coupled by the common-base amplifier action of transistor 14 to compound transistor 20, in turn increasing its conduction. The non-conduction of compound transistor structure 18 and the increased conduction of compound transistor structure 20 causes the output signal at output terminal 21 to exhibit a more positive potential than during quiescent conditions. 
     On the other hand, a negative signal current variation applied to the joined base electrodes of transistors 11 and 13 increases conduction through each of them. The increased conduction of 11 causes an increase in the portion of I 17  supplied to compound transistor 18, increasing its conduction. The increased conduction of transistor 13 causes it to appropriate to itself all of I 19 , rendering transistor 14 and consequently compound transistor structure 20 non-conductive. The result is that the output signal at output terminal 21 exhibits a less positive potential than during quiescent conditions. 
     Compound transistor 18 includes a large transistor 181, which in fact may be several smaller transistors operating in parallel. Transistor 181 is connected in cascade after 182 in a connection much like the conventional Darlington cascade. A third transistor 183 is connected to stabilize the current gain of the cascaded transistor 181 and 182 to predictable values despite their common-emitter forward current gains (h fe  &#39;s) exhibiting the normal range of manufacturing tolerances. 
     Predictable low-level current gains are desired of compound transistors 18 and 20 so their idling current can be set to the lowest level consistent with low cross-over distortion in their Class B push-pull operation, the ultimate aim being to reduce standby thermal dissipation from compound transistors 18 and 20. 
     At low idling current levels, the potential drops across the emitter degeneration resistors 184 and 185 of 181 and 183, respectively, will be negligibly small. The internal emitter resistances of 181 and 183 will be much larger than the resistances of resistors 184 and 185, respectively. Under these circumstances, resistors 184 and 185 have negligible effect upon operation of compound transistor 18. Compound transistor 18 behaves as a current mirror amplifier, its current gain being equal to the transconductance of 181 divided by the transconductance of 183. The transconductances of transistors with similar base-emitter junction doping profiles are related proportionally to the areas of their base-emitter junctions. Transistor 183 is usually made relatively small as compared to 181, so the transconductance of 183 is smaller than that of 181 by a factor. This factor is equal to the low-level current gain of compound transistor 18. 
     Predictable high-level current gains are desired of compound transistors 18 and 20 so they may be afforded a predictable degree of over-current protection by controlling the input current drive applied to their effective base electrodes. 
     At higher current levels, as are encountered during peaks of output signal, the internal emitter resistances of transistors 181 and 183 are no longer high compared to the resistances of resistors 184 and 185. The effective transconductances of transistors 181 and 183 will be reduced by current feedback across their degeneration resistors, and the gain of the &#34;current mirror amplifier&#34; comprised by compound transistor 18 will, for high signal current levels, be the resistance of resistor 185 divided by that of resistor 184. This high-level current gain is usually made much larger than the h fe  of 181, but smaller than the product of the h fe  &#39;s of transistors 181 and 182. 
     Resistor 186 is a pull-down resistor for the emitter electrode of 182. Elements 201, 202, 203, 204, 205, 206 of compound transistor 20&#39; correspond respectively to elements 181, 182, 183, 184, 185, 186 of compound transistor 18&#39;. 
     The controlled current source 17 supplying the joined emitter electrodes of transistors 11 and 12 comprises those elements with reference numerals starting with the two digits 1, 7. A direct potential V A  developed by means not shown, is appreciably smaller than the V B  operating potential applied to terminal 27, but larger than the V BE  offset potential across the base-emitter junction of a transistor --e.g., about 1-2 volts. Transistor 171 is provided with an emitter degeneration resistor 172 and has V A  applied to its base electrode. This conditions transistor 171 to demand a constant collector current I C171  from the input port of a current mirror amplifier 173 comprising transistors 174, 175, 176. The output port of current mirror amplifier 173 supplies the joined emitter electrodes of transistors 11 and 12 with a component of current proportionally related to I C171 . This current component provides a lower limit to the base drive current available to transistor 181 during signal peaks no matter what the emitter-to-collector potential of 181 is. This limit is chosen so the collector current I U , signal peaks may cause to flow in transistor 181, cannot over-dissipate 181 despite full operating potential V B  appearing between its emitter and collector electrodes. Such condition might obtain were output terminal 21 accidentally shorted to terminal 27, for example. 
     The controlled current source 19 includes elements 191, 192, 193, 194, 195 and 196 which operate in the same manner as elements 171, 172, 173, 174, 175 and 176, respectively, of source 17 as thus far described. Just as a small collector current I U  can be caused to flow to transistor 181, no matter what its collector current and emitter-to-collector potential conditions are, for similar reasons a small emitter current I U  &#39; can be caused to flow from transistor 201, no matter what its collector current and emitter-to-collector potential conditions are. This ensures that there will always be some current available from the amplifier to charge or discharge a coupling capacitor (not shown) used to connect output terminal 21 to a low-impedance load (not shown). In an amplifier provided with protection against over-dissipation of its output transistors--e.g., the FIGURE amplifier--redistribution of charge on the coupling capacitor can activate the protection circuitry. An amplifier without specific means for charging on or discharging the coupling capacitor exhibits a protracted period between applying operating potential and having full amplification capability available, which is undesirable. 
     Current mirror amplifier 173 in controlled current source 17 also receives I C177 , the collector current of transistor 177, as input current. I C177  varies as a function of the emitter-to-collector potential, V CE181 , of 181, as is explained hereinafter. At low values of V CE181 , I C177  is at its maximum. This permits the maximum rated collector current of transistor 181 to flow responsive to increased collector current demands of transistors 30 and 31. But, as V CE181  increases beyond a particular threshold potential V T , I C177  is reduced proportional to (V CE181  - V T ), curtailing the base current drive available to 181 substantially proportionally to (V CE181  - V T ), and thereby reducing its maximum collector current commensurately. Finally, at a value of V U  of V CE181 , I C177  becomes zero-valued and the collector current of 181 can have a maximum value only of I U , available in response to the continuing flow of I C177 . 
     FIG. 2 shows this overload limiting characteristic plotted over the collector characteristics of transistor 181. The maximum current flows permitted by overload limiting characteristics are substantially higher than that to be expected from rated minimum resistive load for maximum rated output current I MAX . This accommodates the driving of inductive loads such as loudspeakers. The sloping portion of the overload limiting characteristic is located so as to prevent the transistor from operating in the region where its collector current characteristics fold back--that is, so as to avoid second breakdown. 
     Referring back to FIG. 1, the control of I C177  will now be considered. I C177  is substantially equal to the emitter current flow in 177, which, in turn, is by Ohm&#39;s Law, equal to the potential drop V 178  across resistor 178 divided by the resistance of R 178 . Potential regulator 40 comprises a series-regulator transistor 41 maintaining a seven volt potential between its emitter and collector electrodes responsive to collector-to-base feedback applied by a potential divider comprising avalanche diode 42 and resistor 43. The network comprising resistors 44 and 45 and a potential regulator 40 establishes a potential at the base electrode of 177 about 3.5 volts more positive than V B  /2, the quiescent value of V 21  at output terminal 21. The emitter-follower action of 177 places its emitter electrode at a quiescent potential about 2.8 volts more positive than V B  /2. So, V 178  equals 2.8 volts, and the resistance of resistor 178 can be chosen to obtain the desired value of I C181  (e.g., one-half I MAX ) for the V CE181  = V B  /2 condition. 
     As V 21  swings to a more positive value, increasing V CE181 , V 178  is reduced, commensurately reducing I E177  and thus I C177  proportionately. When V 21  exceeds a V U  equal to V B  /2 plus 2.8 volts, transistor 177 is cut-off and I C177  equals zero. 
     When V 21  swings to a less positive value than V B  /2 the potential drop across 178 increases, increasing I C177  proportionately. As V 21  becomes still less positive, reducing V CE181  to a value V T , the base-emitter junction of 46 is biased into conduction. As V 21  becomes still less positive, this reduction in potential is coupled by the emitter-follower action of transistor 46 and the offset potential across potential regulator 40 to the base electrode of 177. The emitter-follower action of 177 causes a reduction in potential at the emitter electrode of 177 corresponding to that in V 21  for this V 21  &lt; V T  condition. This keeps V 178  substantially constant, causing substantially constant I E177  flow through 178, causing I C177  to be substantially constant despite V 21  becoming increasingly less positive. 
     Transistor 201 is provided similar overload protection to that afforded to transistor 181, by controlling the collector current I C197  of transistor 197 in complementary fashion to I C177 . The network comprising resistors 44 and 45 and potential regulator 40 establishes a potential at the base electrode of 197 about 3.5 volts less positive than V B  /2, the quiescent value of V 21 . The emitter-follower action of 197 places its emitter electrode at a quiescent potential about 2.8 volts less positive than V B  /2 so V 198 , the potential drop across resistor 198, equals 2.8 volts, and the resistance R 198  of resistor 198 can be chosen to obtain a desired value of I C197 . I C197  is applied to the input circuit of a current mirror amplifier 199 having minusunity current gain, and the output current from current mirror amplifier 199 is supplied to the input circuit of current mirror amplifier 193. Current mirror amplifier 193 supplies, therefore, a current to the joined emitter electrodes of transistors 13 and 14 which is proportional to I C197 . 
     As V 21  swings to a less positive value, increasing the emitter-to-collector potential V CE201  of transistor 201, V 198  is reduced commensurately, reducing I C197  proportionately. When V CE201  exceeds V U , 197 is cut-off and I C197  equals zero. When V 21  swings to a more positive value than V B  /2 the potential drop across resistor 198 increases, increasing I C197  proportionately. As V 21  becomes still more positive, reducing V CE201  to a value V T , the base-emitter junction of transistor 47 is biased into conduction. Thereupon, as V 21  becomes still more positive, the change in this potential is coupled by the potential follower actions of transistor 47, potential regulator 40 and transistor 197 to maintain V 198  substantially unchanged. This causes I C197  to remain constant for this V 21  &gt; V B  -V T  condition. 
     Provided with the foregoing description of the over-load protection scheme involved in the present invention, the skilled circuit designer can design modifications to tailor the over-load limiting characteristic to particular designs of output transistors 181, 201. The volts/milliamperes slope of the falling portion of the characteristic is governed by the resistance of resistors 178 and 198. The current at the intercept of this slope with V B  /2 potential is controlled by selection of the quiescent base potentials of 177 and 197. With the connections of transistors 46 and 47 shown in FIG. 1, I MAX  is substantially twice the current at that intercept but offsetting the emitter electrodes of 46 and 47 from the base electrodes of 197 and 177, respectively, can alter this relationship. 
     The over-load protection scheme used in the FIG. 1 circuit invites comparison with Leidich&#39;s. Leidich supplies temperature-compensated base currents to the output transistors of his amplifier, which transistors are free to exhibit temperature dependencies in their current gains. In the FIG. 1 circuit, the current gains of the output transistors 181 and 201 are stabilized to be independent of temperature by their connection in the &#34;current mirror amplifiers&#34; 18 and 20, as previously described; and the base current drives are not inversely related to the current gains of the output transistors nor need they be so related in order to obtain predictable over-load current. This is desirable since the open-loop gain of the amplifier can be kept in constant with temperature, simplifying the designing of the amplifier into feedback loops. Leidich provides the same degree of over-load current limiting for all emitter-to-collector potential conditions on the output transistors of his amplifier, sacrificing the ability to operate over a substantial portion of their safe operating area ratings (SOAR&#39;s). 
     The offset potential developed across the potential regulator comprising elements 32-34 forestalls the possibility of forward-biasing of the collector-to-base junction of transistor 11 when transistor 30 is driven into saturation. 
     Transistor 51 is provided with an emitter degeneration resistor 52 and has bias potential V A  applied to its base electrode. This conditions 51 to withdraw a constant collector current from the input circuit of a current mirror amplifier 53 comprising elements 54, 55, 56 and 36. Amplifier 53 responds to supply a constant output current from the collector electrode of 36. 
     The configuration 60 including elements 61-70 develops the ΔV BE  potential applied between the base electrodes of 12 and 14, and in essence takes the form of a two-terminal current regulator circuit between V 21  and V B . Elements 61-63 form a current amplifier 64 with an input terminal at the base electrode of 61, a common terminal at the joined emitter electrodes of transistors 61 and 62, and an output terminal at the collector electrode of transistor 62. At relatively low current levels, the current gain of amplifier 64 is the ratio of the transconductance of 62 to the transconductance of 61, the potential drop across resistor 63 being negligible so amplifier 64 functions like a conventional simple current mirror amplifier. At higher current levels, the potential drop across resistor 63 reduces the base-emitter potential of transistor 62 vis-a-vis that of transistor 61 and reduces the current gain of amplifier 64. Elements 65-68 form a current amplifier 69 with an input terminal at the connection of the collector electrode of 65 and the base electrode of 67; a common terminal at the emitter electrode of 66; and an output terminal at the collector electrode of 66. At relatively low current levels, the current gain of amplifier 69 is the ratio of the transconductance of 66 to the transconductance of 65, the potential drop across resistor 68 being negligible so amplifier 69 functions like a well-known current mirror amplifier. At higher current levels, the potential drop across resistor 68 increases, so the base-emitter potential of 66 vis-a-vis that of 65 increases, increasing the current gain of current amplifier 69. The output terminal of current amplifier 64 is connected to the input terminal of current amplifier 69; and the output terminal of amplifier 69 is connected via resistor 70 to the input terminal of amplifier 64, thus forming a positive feedback loop. 
     Assume the low-level current gains of current amplifiers 64 and 69 are each made to be minus unity by matching the transconductances of 61 and 62 to each other and by matching the transconductances of 65 and 66 to each other. Then, at relatively low current levels, the current flowing through the collector-emitter paths of 66 and 61 would be substantially the same in amplitude as the current flowing in the collector-emitter paths of 62 and 65. The base currents of 12 and 14 applied to the input circuit of current amplifier 64 give rise to such current flows. Responsive to these flows, the potential drop across resistor 68 will exceed that across resistor 63 because the resistance of the resistor 68 exceeds that of resistor 63. This will tend to increase the current gain of amplifier 69 by a greater factor than the current gain of amplifier 64 is reduced, so the regenerative feedback loop exhibits an open-loop gain greater than unity. In the closed loop, this results in a growth of the current levels in the loop. In this regenerative process, the current flow through the collector-to-emitter paths of 66 and 61 increases more rapidly than that through the collector-emitter paths of 62 and 65. Therefore, the potential drop across resistor 68 tends to more closely approach that across resistor 63 despite both of these potential drops tending to increase. This means the current gain of the amplifier 69, although increasing, will at some point in the regenerative process, begin to increase less rapidly than the gain of current amplifier 64 is decreased. When the potential drops across resistors 68 and 63 become equal, an equilibrium condition is reached where the current gains of amplifiers 64 and 69 are in reciprocal relationship to each other. This causes the positive feedback loop to have an open-loop gain of unity, and there is no further regenerative growth in the currents in the loop. 
     It can be shown, proceeding from a standard equation describing transistor operation, that when this equilibrium condition obtains, the potential drop across resistors 70 and 63 has the value (kT/q) (R 63  +R 70 ) (1/R 63 ) ln (R 68  /R 63 ) in which R 63 , R 68  and R 70  are the respective resistances of resistors 63, 68 and 70. 
     FIG. 3 shows a Class B push-pull amplifier similar to that of FIG. 1 insofar as how load-line limiting is carried out. However, the relative biasing of the base electrodes of transistors 11, 12, 13 and 14 is carried out differently. Transistors 12 and 14 have interconnected base electrodes and a ΔV BE  offset potential drop across resistor 37 biases the base electrode of transistor 11 more positive than that of transistor 13. 
     Configuration 60&#39; resembles configuration 60, but is connected between the operating potential terminals 27 and 28, rather than between terminal 27 and a point closely following the output terminal 21 in potential. Also, current amplifier 69&#39; includes a resistor 70&#39;. Resistors 70&#39; and 68 have resistances R 70  and R 68 , respectively, related in a proportion reciprocally related to the effective base-emitter junction areas of transistors 66 and 65. Accordingly, current amplifier 69&#39; is a current mirror amplifier with fixed current gain, -G, G being less than unity. Equilibrium obtains when the potential drops across resistor 63 equals -(kT/q) ln G. 
     Transistor 36 and its emitter degeneration resistor 38 are proportioned vis-a-vis transistor 66 and its emitter degeneration resistor 70, respectively, so collector current flow from transistor 36 is proportional to the collector current flow of transistor 66, causing the potential drop across resistor 37 to be proportional to that across resistor 63. Transistor 361 and its emitter degeneration resistor 381 are proportioned vis-a-vis transistor 66 and its emitter degeneration resistor 70&#39;, respectively, so collector current flow from transistor 361 is proportional to the collector current flow of transistor 66, causing the potential drop across resistor 371 to be proportional to that across resistor 63. The FIG. 3 configuration is advantageous over that of FIG. 1 in eliminating need for elements 51-56. 
     Amplifiers are possible, similar to those in FIGS. 1 and 3, but in which the serial connection of compound transistor structures 18 and 20 is reversed, the application of overall feedback is appropriately modified to maintain it degenerative, I 19  is applied to the joined emitter electrodes of transistors 11 and 12, I 17  is applied to the joined emitter electrodes of transistors 13 and 14, and ΔV BE  biasing is applied between the base electrodes of transistors 12 and 14. Such amplifiers embody the present invention. 
     A number of other embodiments of the present invention will occur to the skilled electronic design engineer and this should be considered in construing the scope of the following claims.