Temperature compensated switching circuit
The present invention is directed to a switching circuit for controlling an output level through a bipolar transistor having a substantially constant saturation voltage thereacross as temperature changes. Many approaches are known in the circuitry arts to the general problem of temperature compensation for bipolar transistors. However, all of these approaches suffer from the effects of saturation as temperature rises or voltage breakdown where antisaturation measures are incorporated and therefore, leave something to be desired when utilized in switching circuits. 
     In the switching circuit of the present invention, an output level is controlled through a bipolar transistor which has the saturation voltage thereacross held substantially constant by means for varying the base current of the bipolar transistor inversely with temperature changes. High temperature nonsaturating means for interrupting the base current to the bipolar transistor may also be included in certain embodiments of the invention to thereby enhance the turn-on speed of the bipolar transistor. 
    
    
     In the drawings: 
     FIG. 1 is a schematic-block diagram of a pulse amplifier having the switching circuit of the invention disclosed therein along with a second switching circuit to respectively control the low and high output levels thereof; 
     FIG. 2 is a schematic diagram for one circuit embodiment of the invention; and 
     FIG. 3 is a schematic diagram for another circuit embodiment of the invention. 
    
    
     Turning now to the drawings, a pulse amplifier 10 having the switching circuit 12 of this invention incorporated therein is shown in FIG. 1. Amplifier 10 includes an input terminal 14 and an output terminal 16 between which the switching circuit 12 is disposed to control whether terminal 16 receives a low voltage level from a source terminal V L . In the switching circuit 12, the main conduction path of a bipolar transistor Q 1  is connected between terminals 16 and V L , while terminal 14 is connected to the base of Q 1  through a compensation means 18 for varying the flow of base current into Q 1  to hold the saturation voltage thereof substantially constant as temperature changes. It should be understood without further explanation that compensation means 18 and Q 1  are thermally coupled to have continuously related temperatures. A second switching circuit 20 is disposed to control whether terminal 16 receives a high voltage level from a source terminal V H . In switching circuit 20, the drain-source conduction path of a MOS/FET transistor Q 2  is connected between terminals 16 and V H , while terminal 14 is connected to the gate of Q 2  through a compensation means 22 for varying the gate-source voltage of Q 2  to hold the drain-source channel resistance thereof substantially constant as temperature changes. As is conventional in pulse amplifiers, a control signal is applied at terminal 14 to render switching circuits 12 and 20 separately conductive to provide either the low or high voltage level at terminal 16. Switching circuit 20 is disclosed and claimed in my copending application (RCA 71,900) that was filed concurrently with this application and is incorporated by reference thereinto. 
     A load would be connected at terminal 16 and affects the voltage level thereat when either switching circuit 12 or 20 is rendered conductive by the control signal at terminal 14. However, compensation means 18 in switching circuit 12 operates to hold the saturation voltage drop across Q 1  substantially constant as temperature changes. Therefore, the current and voltage available for the load at terminal 16 will be substantially constant, if it is assumed that the effect of the load is properly temperature compensated. 
     Switching circuit 12 is useful in applications other than the pulse amplifer 10 where low drift outputs are important as temperature changes. In applications where the load on the switching circuit 12 is not temperature compensated, the saturation voltage drop across Q 1  can be controlled with the compensation means 18 to provide such temperature compensation. Furthermore, a switching circuit similar to switching circuit 12 but with a transistor of complementary conductivity type to that of Q 1  could replace switching circuit 20 in the pulse amplifier 10 of FIG. 1. 
     A more specific embodiment of the switching circuit 12&#39; is illustrated in FIG. 2 where a Schottky transistor Q 3  and a resistor R 1  are disposed in the compensation means 18&#39; thereof. Within Q 3 , a Schottky barrier diode (not shown separately) is connected between the collector and the base of a bipolar transistor to provide a parallel conduction path across the collector-base junction thereof. When the transistor is conductive, its collector-base junction is clamped at the forward-bias voltage of the diode and therefore, the switching delay caused by saturation of the transistor is minimized. R 1  is connected between terminal V&#39; H  and one side of main conduction path through Q 3 , while terminal V&#39; L  is connected to the other side of the main conduction path through Q 3 . Terminal 14&#39; is at the base of Q 3  and is the signal input for switching circuit 12&#39; at which the control signal is applied. Q&#39; 1  is connected with its base to the interconnection between R 1  and the main conduction path of Q 3 , and with its main conduction path between terminal 16&#39;  and V&#39; L . The signal output from switching circuit 12&#39; is taken at terminal 16&#39;. 
     When a low level control signal is applied at terminal 14&#39;, Q 3  is rendered nonconductive to in turn render Q&#39; 1  conductive, and switching circuit 12&#39; operates to apply a voltage at terminal 16&#39;. Because the saturation voltage of Q&#39; 1  tends to vary directly with temperature, R 1  is selected to have a positive temperature coefficient so that the base current of Q&#39; 1  will vary inversely with temperature. Therefore, by scaling the resistance value and the temperature coefficient of R 1  in accordance with the saturation voltage temperature coefficient of Q&#39; 1 , the saturation voltage drop across Q&#39; 1  is held substantially constant as temperature changes. 
     Although the nonsaturation characteristics of the Schottky transistor Q 3  in the compensation means 18&#39; of FIG. 2 are quite satisfactory for many applications, in high temperature environments such characteristics deteriorate. Use of the Schottky transistor can be avoided in another embodiment of switching circuit 12&#34; which is illustrated in FIG. 3. In the compensation means 18&#34;, R&#39; 1  is connected between terminal V&#34; H , and V&#34; L  through the main conduction path of a bipolar transistor Q 4 . The base of Q 4  is connected to terminal V&#34; L  through a resistor R 2  and to terminal V&#34; H  through the main conduction path of another bipolar transistor Q 5  in series with a resistor R 3 . The base of Q 5  is commonly connected to one electrode of separate diodes D 1  and D 2  which are each derived from one junction of a bipolar transistor having its other junction shunted. This common connection is also connected to terminal V&#34; H  through a resistor R 4 . Terminal 14&#34; is at the other electrode of D 1  and is the signal input for switching circuit 12&#34; at which the control signal is applied. The other electrode of D 2  is connected through a resistor R 5  to the interconnection between R&#39; 1  and the main conduction path of Q 4 , while Q&#34; 1  is connected with its base to that same interconnection and with its main conduction path between terminals 16&#34; and V&#34;.sub. L. The signal output from switching circuit 12&#34; is taken at terminal 16&#34;. 
     When a high level control signal is applied at terminal 14&#34; of sufficient magnitude to reverse bias D 1 , the base-emitter junction of Q 5  becomes forward-biased and Q 5  draws base current from V&#34; H  through R 4  to rapidly become conductive. The resistance values of R 2  and R 3  are selected to establish a voltage drop across R 2  of sufficient magnitude to forward-bias the base-emitter junction of Q 4  which also becomes conductive rapidly. The main conduction path of Q 4  then draws current from V&#34; H  through R&#39; 1 . Initially Q 4  will go into saturation shunting the base of Q&#34; 1  with a low resistance path to V&#34; L  causing rapid turn off of Q&#34; 1 . Thereafter, the voltage on the base of Q&#34; 1  reaches a level sufficient to forward-bias the junction of D 2  relative to the voltage on the base of Q 5 , and base current is diverted from Q 5  through R 5  to pull Q 4  out of saturation. The voltage level on the base of Q 5  is above V&#34; L  by a magnitude equal to V BEQ4  +V BEQ5  so that D 2  remains nonconductive until the voltage on the base of Q&#34; 1  drops below this voltage level by at least the barrier voltage of its junction. 
     With Q 4 , Q 5  and D 2  all conductive, the collector current of Q 4  is drawn from V&#34; H  through R&#39; 1  and through R 5 , D 2  and R 4 . Consequently, Q 4  is provided with a collector-to-base feedback connection through R 5 , D 2  and the emitter-follower action of Q 5  to relate both the base current of Q 5  and the collector current of Q 4  to the voltage drop across R 4 . This voltage drop stabilizes at a value equal to V&#34; H  -(V BEQ5  +V BEQ4 ). Since Q 4  must rapidly become nonconductive if Q&#34; 1  is to become rapidly conductive, R 5  is sized to hold the voltage on the base of Q&#34; 1  below its V BE  level, while holding the collector-emitter voltage across Q 4  above its saturation level. Assuming that the current through R 5  is substantially equal to the current through R 4 , and the junction barrier voltage of D 2  and the V BE  of Q 4  and Q 5  are substantially identical, such as would be the case in an integrated circuit, this is accomplised in the circuitry of FIG. 3 by making the value of R 5  =R 4  (V.sub. BE/2)/(V&#34; H  -2 V BE ). 
     When a low level control signal is applied at terminal 14&#34; to forward-bias D 1 , both the base-emitter junction of Q 5  and the junction of D 2  become reverse-biased. Q 5  is rendered nonconductive to in turn render Q 4  nonconductive, while D 2  is rendered nonconductive to discontinue current flow through R 5 . Because Q 4  had previously been in nonsaturated conduction, it turns off rapidly to forward bias the base-emitter junction of Q&#34; 1  which turns on rapidly to draw base current from V&#34; H  through R&#39; 1 . Because the saturation voltage of Q&#34; 1  tends to vary directly with temperature, R&#39; 1  is selected to have a positive temperature coefficient so that the base current of Q&#34; 1  will vary inversely with temperature. Therefore, by selecting the value and temperature coefficient for R&#39; 1  in accordance with the saturation voltage temperature coefficient of Q&#34; 1 , the saturation voltage drop across Q&#34; 1  is held substantially constant as temperature changes. 
     Those skilled in the art should understand without further explanation that the high voltage source terminal rather than the low voltage source terminal could have been connected to the output terminal through the switching circuit of the invention in the arrangements of FIGS. 1, 2 and 3. The transistors in such switching circuits would be of the PNP conductivity type, rather than the NPN conductivity type which are shown. Therefore, this invention has been disclosed herein by describing only selected embodiments thereof and it should be understood by those skilled in the art that numerous changes in the details of construction and the combination or arrangement of parts could be made in the described embodiments without departure from the true scope or spirit of the invention. Consequently, the present disclosure should be construed as illustrative rather than limited.