Temperature compensated switching circuit
The present invention is directed to a switching circuit for controlling an output level through a MOS/FET transistor having a substantially constant channel resistance thereacross as temperature changes. Many approaches are known in the circuitry arts to the general problem of temperature compensation. In the art of integrated circuitry, however, most known approaches apply only to circuits having bipolar transistors and when temperature compensated, these circuits either demonstrate a slow switching speed or encounter voltage breakdown at high output levels. 
     In the switching circuit of the present invention, an output level is controlled through a MOS/FET transistor and means for varying the gate-source voltage of this transistor directly with temperature is included to hold the channel resistance thereof substantially constant as temperature changes. This switching circuit is particularly appropriate for use in a pulse amplifier to control each output level thereof. It can also be utilized to control one output level in the pulse amplifier, while a second switching circuit controls the other output level thereof through a bipolar transistor. Embodiments utilizing this second switching circuit may include means for varying the base current of the bipolar transistor inversely with temperature to hold the saturation voltage thereacross substantially constant as temperature changes. HIgh temperature nonsaturating means for interrupting the base current to the bipolar transistor may also be included in those embodiments where the second switching circuit is utilized, to enhance the turn-on speed of the bipolar transistor. 
    
    
     In the drawings: 
     FIG. 1 is a schematic-block diagram of a pulse amplifier having the switching circuit of the invention disposed therein along with a second switching circuit to respectively control the high and low output levels thereof; 
     FIG. 2 is a schematic diagram for one circuit embodiment of the pulse amplifier shown in FIG. 1; and 
     FIG. 3 is a schematic diagram for another embodiment of the second switching circuit in the pulse amplifier of FIG. 1. 
    
    
     Turning now to the drawings, a pulse amplifier 10 having the switching circuit 12 of this invention incorporated threin is shown in FIG. 1. A signal applied to the input terminal 14 of amplifier 10 controls whether the output terminal 16 receives a high voltage from the source terminal V H  or a low voltage from the source terminal V L . The switching circuit 12 selectively connects terminal V H  to terminal 16 through the drain-source conduction path of a MOS/FET transistor Q 1 . Terminal 14 is connected to the gate of Q 1  through a compensation means 18 for varying the gate-source voltage of Q 1  to hold the drain-source channel resistance thereof substantially constant as temperature changes. A second switching circuit 20 connects terminal V L  to terminal 16 through the main conduction path of a bipolar transistor Q 2 . Input terminal 14 is connected to the base of Q 2  through a compensation means 22 for varying the flow of base current into Q 2  to hold the saturation voltage thereof substantially constant as temperature changes. It should be understood without further explanation that compensation means 18 is thermally coupled with Q 1  and compensation means 22 is thermally coupled with Q 2  so that continuously related temperatures exist therebetween. 
     A load would be connected at terminal 16 and affects the voltage level thereat when either switching circuit 12 or 20 is rendered conductive by the control signal at terminal 14. However, compensation means 18 operates to hold the channel resistance of Q 1  substantially constant as temperature changes, while compensation means 22 operates to hold the saturation voltage across Q 2  substantially constant as temperature changes. Therefore, the current and voltage available for the load at terminal 16 will be substantially constant whether switching circuit 12 or 20 is rendered conductive by the control signal, if it is assumed that the effect of the load is properly temperature compensated. As is conventional in pulse amplifiers, the control signal at the input terminal 14 renders only switching circuit 12 or switching circuit 20 conductive at any given time. 
     The switching circuit 12 of this invention is useful in applications other than the pulse amplifier 10 where low drift outputs are important as temperature change. In applications where the load on the switching circuit 12 is not temperature compensated, the channel resistance of the MOS/FET transistor can be controlled to provide such temperature compensation. Futhermore, switching circuit 12 with a MOS/FET transistor of complementary conductivity type to that of Q 1  could replace switching circuit 20 in amplifier 10 of FIG. 1. 
     A specific circuit embodiment for the amplifier 10 of FIG. 1 is illustrated in FIG. 2. In the switching circuit 12&#39; of this embodiment, the gate of the MOS/FET transistor Q&#39; 1  is separately connected to the signal output from a nonsaturating voltage comparator 24 through a diode D 1  and to the source terminal V&#39; H  through the main conduction path of a bipolar transistor Q 3 . D 1  is poled for conduction therethrough to increase the gate-source voltage of Q&#39; 1  in accordance with enhanced drain-source channel conductivity thereof. The base of Q 3  is connected to the signal output from comparator 24 which renders Q 3  conductive to charge a gate/source capacitance whenever D 1  is reversed-biased thereby. Bipolar transistors Q 4  and Q 5  are differentially connected in comparator 24 with one side of their main conduction path commonly connected to one side of the main conduction path through a bipolar transistor Q 6 . The other sides of the main conduction paths through Q 4  and Q 5  are connected to terminal V&#39; H  through resistors R 1  and R 2 , respectively. The other side of the main conduction path through Q 6  is connected to terminal V&#39; L  through a resistor R 3 . Resistors R 4 , R 5  and R 6  are successively connected in series between terminals V&#39; H  and V&#39; L  with the base of Q 4  being connected at the interconnection between R 4  and R 5 , while the base of Q 6  is connected at the interconnection between R 5  and R 6 . The control signal on terminal 14&#39; of amplifier 10&#39; is connected to the signal input of comparator 24 at the base of Q 5 . 
     When the control signal on terminal 14&#39; exceeds the level that is applied at the base of Q 4 , Q 5  becomes conductive to reduce the signal output level from comparator 24 relative to terminal V&#39; H . This lower signal level causes current to flow through D 1 , which establishes a gate-source voltage differential within Q&#39; 1  of a polarity to render the drain-source channel thereof conductive. If the gate-source voltage of Q&#39; 1  were fixed, the channel resistance thereof would tend to vary directly with temperature, so that the voltage drop thereacross and the current flow therethrough would not be held constant for a given load. Therefore, Q 6  is disposed in comparator 24 to regulate current flow through Q 5  and thereby vary the gate-source voltage of Q&#39; 1  to hold the drain-source channel resistance thereof substantially constant as temperature changes. Because Q 6  is connected in a common-base configuration, its collector current will be substantially equal to its emitter-current. Furthermore, both the collector current and emitter current of Q 6  will demonstrate a positive temperature coefficient due to the negative temperature coefficient that is exhibited by the base-emitter junction resistance of Q 6  and the constant voltage that is applied to the base of Q 6  by the voltage divider network of R 4 , R 5  and R 6  as temperature changes. Of course, R 3  will affect the value of emitter current drawn by Q 6  while both R 2  and R 3  will affect the gate-source voltage of Q 1  that is derived from collector current flowing through Q 6 . Consequently, by selecting R 2  and R 3  with compatible positive temperature-coefficients to the negative temperature-coefficient for the base-emitter junction resistance of Q 6 , the gate-source voltage of Q 1  will demonstrate the positive temperature-coefficient that is necessary to hold the drain-source channel resistance of Q 1  substantially constant as temperature changes. Furthermore, saturated conduction through Q 5  in comparator 24 can be precluded to enhance the speed at which Q 1  is rendered nonconductive. This is accomplished by selecting the values of R 4 , R 5  and R 6  so that the base of Q 6  is biased to limit current flow through the main conduction path of Q 5  to a level below that required thereby for saturation. 
     A Schottky transistor Q 7  and a resistor R 7  are disposed within the compensation means 22&#39; of the switching circuit 20&#39; in FIG. 2. Within Q 7 , a Schottky barrier diode (not shown separately) is connected between the collector and the base of a bipolar transistor to provide a parallel conduction path across the collector-base junction thereof. When the transistor is conductive, its collector-base junction is clamped at the forward-bias voltage of the diode and therefore, the switching delay caused by saturation of the transistor is minimized. R 7  is connected between terminal V&#39; H  and one side of main conduction path through Q 7 , while terminal V&#39; L  is connected to the other side of the main conduction path through Q 7 . Terminal 14&#39; is at the base of Q 7  and is the signal input for switching circuit 20&#39; at which the control signal is applied. Q&#39; 2  is connected with its base to the interconnection between R 7  and the main conduction path of Q 7 , and with its main conduction path between terminals 16&#39; and V&#39; L . The signal output from switching circuit 20&#39; is taken at terminal 16&#39;. Resistor R 12  limits the base current of Q 7 . 
     When a low level control signal is applied at terminal 14&#39;, Q 7  is rendered nonconductive to in turn render Q&#39; 2  conductive, and switching means 20&#39; operates to apply a voltage at terminal 16&#39;. Because the saturation voltage of Q&#39; 2  tends to vary directly with temperature, R 7  is selected to have a positive temperature coefficient so that the base current of Q&#39; 2  will vary inversely with temperature. Therefore, by scaling the resistance value and the temperature coefficient of R 7  in accordance with the saturation voltage temperature coefficient of Q&#39; 2 , the saturation voltage drop across Q&#39; 2  is held substantially constant as temperature changes. 
     Although the nonsaturation characteristics of the Schottky transistor Q 7  in the compensation means 22&#39; of FIG. 2 are quite satisfactory for many applications, in high temperature environments such characteristics deteriorate. Use of the Schottky transistor can be avoided in another embodiment of switching circuits 20&#34; which is illustrated in FIG. 3. In the compensation means 22&#34;, R&#39; 7  is connected between terminal V&#34; H , and V&#34; L  through the main conduction path of a bipolar transistor Q 8 . The base of Q 8  is connected to terminal V&#34; L  through a resistor R 8  and to terminal V&#34; H  through the main conduction path of another bipolar transistor Q 9  in series with a resistor R 9 . The base of Q 9  is commonly connected to one electrode of separate diodes D 2  and D 3  which are each derived from one junction of a bipolar transistor having its other junction shunted. This common connection is also connected to terminal V&#34; H  through a resistor R 10 . Terminal 14&#34; is at the other electrode of D 2  and is the signal input for switching circuit 20&#34; at which the control signal is applied. The other electrode of D 3  is connected through a resistor R 11  to the interconnection between R&#39; 7  and the main conduction path of Q 8 , while Q&#34; 2  is connected with its base to that same interconnection and with its main conduction path between terminals 16&#34; and V&#34; L . The signal output from switching circuit 20&#34; is taken at terminal 16&#34;. 
     When a high level control signal is applied at terminal 14&#34; of sufficient magnitude to reverse bias D 2 , the base-emitter junction of Q 9  becomes forward-biased and Q 9  draws base current from V&#34; H  through R 10  to rapidly become conductive. The resistance values of R 8  and R 9  are selected to establish a voltage drop across R 8  of sufficient magnitude to forward-bias the base-emitter junction of Q 8  which also becomes conductive rapidly. The main conduction path of Q 8  then draws current from V&#34; H  through R&#39; 7  to drop the voltage on the base of Q&#34; 2  which becomes nonconductive when its base-emitter junction becomes reverse-biased. When the voltage on the base of Q&#34; 2  drops sufficiently to forward-bias the junction of D 3  relative to the voltage on the base of Q 9 , base current is diverted from Q 9  through R 11 . The voltage level on the base of Q 9  is above V&#34; L  by a magnitude equal to V BEQ8   +V BEQ9  so that D 3  remains nonconductive until the voltage on the base of Q&#34; 2  drops below this voltage level by at least the barrier voltage of its junction. 
     With Q 8 , Q 9  and D 3  all conductive, the collector current of Q 8  is drawn from V&#34; H  through R&#39; 7  and through R 11 , D 3  and R 10 . Consequently, Q 8  is provided with a collector-to-base feedback connection through R 11 , D 3  and the emitter-follower action of Q 9  to relate both the base current of Q 9  and the collector current of Q 8  to the voltage drop across R 10 . This voltage drop stabilizes at a value equal to V&#34; H  -(V BEQ9  +V BEQ8 ). Since Q 8  must rapidly become nonconductive if Q&#34; 2  is to become rapidly conductive, R 11  is sized to hold the voltage on the base of Q&#34; 2  below its V BE   level, while holding the collector-emitter voltage across Q 8  to less than its saturation level. Assuming that the current through R 11  is substantially equal to the current through R 10  and the junction barrier voltage of D 3  and the V BE  of Q 8  and Q 9  are substantially identical, such as would be the case in an integrated circuit, this is accomplished in the circuitry of FIG. 3 by making the value of R 11  =R 10  (V BE/2 )/(V&#34; H  -2V BE ). 
     When a low level control signal is applied at terminal 14&#34; to forward-bias D 2 , both the base-emitter junction of Q 9  and the junction of D 3  become reverse-biased. Q 9  is rendered nonconductive to in turn render Q 8  nonconductive, while D 3  is rendered nonconductive to discontinue current flow through R 11 . Because Q 8  had previously been in nonsaturated conduction, it turns off rapidly to forward bias the base-emitter junction of Q&#34; 2  which turns on rapidly to draw base current from V&#34; H  through R&#39; 7 . Because the saturation voltage of Q&#34; 2  tends to vary directly with temperature, R&#39; 7  is selected to have a positive temperature coefficient so that the base current of Q&#34; 2  will vary inversely with temperature. Therefore, by selecting the value and temperature coefficient for R&#39; 7  in accordance with the saturation voltage temperature coefficient of Q&#34; 2 , the saturation voltage drop across Q&#34; 2  is held substantially constant as temperature changes. 
     Those skilled in the art should understand without further explanation that the high voltage source terminal rather than the low voltage source terminal could have been connected to the output terminal through the switching circuit 20, 20&#39;, and 20&#34; of FIGS. 1, 2 and 3 respectively. Furthermore, the transistors in these switching circuits could also be of the PNP conductivity type, even though only NPN conductivity type are shown. 
     Therefore, this invention has been disclosed herein by describing only selected emobodiments thereof and it should be understood by those skilled in the art that numerous changes in the details of construction and the combination or arrangement of parts could be made in the described embodiments without departure from the true scope or spirit of the invention. Consequently, the present disclosure should be construed as illustrative rather than limited.