Protective circuitry for transistorized d-c/d-c converter
FIELD OF THE INVENTION 
     My present invention relates to a d-c/d-c converter of the type wherein two switching transistors energized by a direct-current source, constituting a d-c/a-c inverter, are operated in push-pull to conduct alternately in response to unbalance currents traversing the primary winding of a self-saturating feedback transformer which has secondary-winding halves inserted in the respective base/emitter circuits of these transistors. 
     BACKGROUND OF THE INVENTION 
     In a conventional converter of this description, the primary of the feedback transformer is connected in series with a resistance across the collector leads of the two switching transistors between which the primary of an output transformer is inserted, the latter primary having a center tap connected to one terminal of the d-c source whose other terminal is connected to the two emitters. The secondary of the output transformer, also split into two halves, works into a full-wave rectifier which includes a shunt capacitance and preferably a series inductance for smoothing purposes. The load connected to this rectifier may also be of a partly capacitive nature. Such a converter has the advantages of high voltage stability with variable loads as well as compactness and high operating efficiency. One drawback, however, resides in its tendency to draw a very large starting current, which could be on the order of five to ten times its steady-state current, on account of the capacitances present downstream of its output transformer. Thus, the switching transistors must be designed to handle such large currents even though that need exists only for a very short time, i.e. in a transient phase following cut-in. 
     A conventional solution to this problem, designed to eliminate the need for oversize transistors, consists in the temporary insertion of a large series resistance into the supply circuit of the converter. Since the protective resistance must be removed or short-circuited after the cut-in but has to be reinserted upon every interruption of normal operation, this solution requires either the assistance of an operator or a complex switching circuit. Such a circuit may comprise, for example, a transistor which acts as the protective resistance by operating linearly in the transient phase but becoming saturated (or being short-circuited) in the steady-state mode. 
     OBJECTS OF THE INVENTION 
     An object of my present invention, therefore, is to provide simple circuitry for limiting the collector current of the switching transistors during start-up of the converter. 
     A related object is to protect these transistors against overloads possibly occurring in operation, e.g. upon an accidental short-circuiting of the load. 
     SUMMARY OF THE INVENTION 
     I have found, in accordance with the present invention, that both these objects can be achieved by the provision of a protective network connected across the primary winding of the feedback transformer to limit the magnitude of the unbalance currents traversing this winding. 
     Pursuant to a more particular feature of my invention, the protective network comprises an ancillary capacitor chargeable through rectifying means such as a diode bridge, this capacitor being provided with a resistive discharge path so as to have zero charge at the time of cut-in. The capacitor, therefore, forms an effective shunt for the feedback primary in the initial phase of operation, i.e. until it has been charged sufficiently to let the converter operate normally. The smoothing and load capacitances downstream of the output transformer are being charged at the same time so that the buildup of the collector currents occurs gradually and without large-amplitude transients. 
     A sudden rise in the collector current due to a short-circuited load, for example, would also bring the protective capacitor into play so as to limit the amplitude of the inverter oscillations. For this latter purpose, however, I may also provide short-circuiting means supplementing or replacing the ancillary capacitor in the protective network, the short-circuiting means being advantageously in the form of at least one thyristor or controlled rectifier and being activable in response to an excessive collector current by current-sensing means such as ancillary transformers having primary windings in series with that of the output transformer. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     The above and other features of the invention will now be described in detail with reference to the accompanying drawing in which: 
     FIG. 1 is a circuit diagram of a transistorized d-c/d-c converter embodying my present improvement; 
     FIGS. 2, 3A and 3B are graphs relating to the operation of the converter shown in FIG. 1; 
     FIGS. 4 and 5 are circuit diagrams representing other embodiments; and 
     FIG. 6 is a circuit diagram relating to a modification of the two latter embodiments. 
    
    
     SPECIFIC DESCRIPTION 
     In FIG. 1 I have shown a d-c/d-c converter comprising a pair of NPN switching transistors T 1  and T 2  with interconnected emitters and with collectors connected to opposite ends of a split primary PP 1 , PP 2  of an output transformer TP. A preferably stabilized source of direct current, generating a supply voltage V a  of 12 or 24 V, for example, has a positive terminal J 1  connected via a switch S to the center tap of transformer winding PP 1 , PP 2  and has a negative terminal J 2  (which could be grounded) connected to the emitters of the two switching transistors. The secondary of output transformer TP is split into two halves SP 1 , SP 2  which are connected via respective diodes D 1  and D 2  as well as a common series inductance L to a positive output terminal O 1 , the corresponding negative terminal O 2  being tied to the junction of winding halves SP 1  and SP 2 . Terminals O 1  and O 2  are connected across a load here schematically represented by a resistance R L  in parallel with a capacitance C L . Inductance L forms part of a low-pass filter including a smoothing capacitance C u  also bridging the output terminals O 1  and O 2 . 
     A self-saturating feedback transformer TA has a primary PA connected in series with a resistor R t  across the collectors of transistors T 1  and T 2  : the magnitude of the unsymmetrically disposed resistor R t  may substantially equal that of the inductance of winding PA at the switching frequency of the d-c/a-c inverter constituted by these transistors. The secondary of feedback transformer TA comprises two winding halves SA 1  and SA 2  respectively connected to the bases of transistors T 1  and T 2  by way of resistors R 1  and R 2  ; other resistors R 11  and R 12  connect these bases to the common emitter lead. The junction of winding halves SA 1  and SA 2  is connected to the center tap of primary PP 1 , PP 2  by way of a start circuit A, here shown to consist simply of a resistor R A , and is further connected to the common emitter lead via the reverse resistance of a diode D 3  preventing the flow of direct current bypassing the windings SA 1  and SA 2  . An input capacitor C a  is connected across source terminals J 1  and J 2 . 
     When the converter so far described is turned on by closure of switch S, the supply current passing from terminal J 1  via resistor R A  and the two circuit branches SA 1 , R 1 , R 11  and SA 2 , R 2 , R 12  to terminal J 2  is not exactly divided between these imperfectly symmetrical branches so that one of the switching transistors, e.g. T 1 , becomes more conductive than the other. The difference in the base voltages of the two transistors is intensified by an unbalance current traversing the primary PA as a result of the different collector potentials. Thus, as is well known per se, the collector current drawn by transistor T 1  through the associated primary half PP 1  progressively increases until feedback transformer TA saturates and causes an increased conductivity of transistor T 2  in series with primary half PP 2  while the current through transistor T 1  diminishes; the switchover between the two transistors continues at a rate determined by the circuit parameters. 
     At the instant of closure of switch S, capacitors C u  and C L  connected across winding halves SP 1  and SP 2  of output transformer TP are discharged and provide a low-impedance path for current pulses induced in the transformer secondary. Inductance L should not be very large in order to avoid excessive lags in the energization and de-energization of the load; it is, therefore, relatively ineffectual in limiting these current pulses which bypass the load resistance R L  and, in the absence of a protective network as described hereinafter, would give rise to objectionably large collector currents. The envelope of such a collector current I c , plotted against time t, has been represented in FIG. 2 by a curve a with an initial peak rising high above the steady-state level I r . 
     The protective network shown in FIG. 1 comprises a diode bridge P which has its input diagonal connected across primary PA and works into a capacitor C p  in parallel with a discharge resistor R p . The magnitude of resistor R p  should substantially exceed that of the resistor R t  in series with winding PA. At the beginning of operations, with capacitors C u , C L  and C p  all discharged, winding PA is virtually short-circuited by this network so as to minimize the regenerative feedback between the collector current and the base current of the more highly conductive transistor. With suitable dimensioning of capacitor C p , taking into account the gain of transistors T 1  and T 2 , there occurs a more gradual buildup of the collector current as represented by a curve b in FIG. 2, that curve rising but slightly above the normal level I r  before merging into same. The transistors, in fact, operate at this stage on a linear part of their characteristics, well below the saturation point. This has been illustrated in FIG. 3A where the collector current I c  of one transistor is shown to start out as a series of pulses of progressively increasing amplitude and relatively high frequency determined in part by the cross-coupling of winding halves SA 1  and SA 2  via the core of feedback transformer TA; at an intermediate stage, at which the protective capacitor C p  is substantially fully charged while the output capacitance C u , C L  may still draw a slight current, these pulses reach a maximum amplitude and width before assuming their steady-state configuration. FIG. 3B shows the corresponding changes of the collector/emitter voltage V ce  of the same transistor which varies in pulses of similar width about the median value V a  and eventually, in steady-state operation, alternates between 2V a  and zero. 
     In FIG. 4 I have illustrated part of the converter of FIG. 1 together with a modified protective network comprising, in addition to ancillary capacitor C p  and its discharge resistor R p , a thyristor or controlled rectifier D p  having its anode/cathode path connected in parallel therewith across the output diagonal of diode bridge P. The gate of thyristor D p  is connected to its cathode on the one hand through a resistor R s  and on the other hand through a pair of transformer windings SB 1  and SB 2  in series with respective diodes D 11  and D 12 . Windings SB 1  and SB 2  are the secondaries of two ancillary transformers TB 1  and TB 2  whose primaries PB 1  and PB 2  are respectively inserted in the collector leads of transistors T 1  and T 2  to act as current sensors. A positive voltage s delivered by secondaries SB 1  and SB 2  to the gate of thyristor D p  is proportional to the magnitude of the collector current and serves as a firing signal when that magnitude exceeds a predetermined threshold, thereby effectively short-circuiting the primary PA of feedback transformer TA. When the collector current has abated sufficiently to quench the thyristor D p , normal operation can resume. 
     The converter of FIG. 4 also comprises a modified start circuit A&#39; including a capacitor C A  in series with resistor R A . This capacitor is designed to prevent the establishment of an anomalous condition in which, with feedback transformer TA virtually deactivated for an extended period as described above, transistors T 1  and T 2  conduct simultaneously with wasteful dissipation of energy. 
     The converter of FIG. 5 differs from that of FIG. 4 in that two thyristors D p  &#39; and D p  &#34; are connected directly and in antiparallel fashion across winding AP, each thyristor being provided with a separate gate resistor R s  &#39; and R s  &#34; respectively connected across the series combination of winding SB 1  with diode D 11  and winding SB 2  with diode D 12 . This protective network operates essentially in the same manner as that of FIG. 4. 
     FIG. 6 shows a further start circuit A&#34; in which the blocking capacitor C A  lies between two resistors R 3 , R 4  in parallel with an NPN transistor T 3  having a collector resistor R 5  and an emitter resistor R 6  ; the base of transistor T 3  is tied to the junction of capacitor C A  with resistor R 4 . On start-up, a positive pulse traversing the capacitor C A  turns on the transistor T 3  to energize the bases of the switching transistors with a rectangular pulse of sufficient magnitude even when capacitor C A  is of small size. The modified start circuit A&#34; of FIG. 6 could be used in lieu of circuit A&#39; of FIGS. 4 and 5; either of these circuits, designed to insulate the transistor bases from the current supply during steady-state operation, can also be employed in the converter of FIG. 1. 
     In lieu of breaking the supply circuit at switch S, an operator could apply an external firing signal s to thyristor D p  (FIG. 4) or thyristors D p  &#39;, D p  &#34; (FIG. 5) to cut off the converter.