Adaptable filtering device and method for filtering sonar signals to remove reverberation signals due to vehicle speed and trim and the sea surface and sea bottom
BACKGROUND OF THE INVENTION 
     The present invention concerns an adaptable filtering device and method for signals received by an active sonar, mounted on a mobile carrier, for rejecting reverberation signals. 
     It is known that a sonar-type apparatus on a carrier vehicle in motion (boat, submarine, torpedo, etc.) detects and localizes other vehicles or objects by emitting ultrasonic waves throughout the surrounding under water space(hereinafter called insonified space) and by observing the echoes received as to their distance and their direction. It is also well known that the ultrasonic energy thus transmitted in the underwater medium is reverberated by particles, bubbles and other discontinuities present in the medium or on its surface. 
     This reverberation interferes with the reception of the echoes since it tends to mask them. One means used to diminish the reverberation is to form at the reception, very directive angular paths, but this means is limited by the dimensions of the antenna. 
     The carrier vehicle being in motion, the frequency of the emitted signal sent back by an immobile target is shifted by the Doppler effect, peculiar to the carrier; this is the case for the reverberation, in which heterogeneities of the medium behave like fixed targets. When the target is in motion, which is the most frequent case, the frequency of the signal sent back by the target is shifted both by the Doppler effect due to movement the carrier and the target. It follows from this that the frequency of the signal corresponding to the reverberation and that corresponding to the target are different. It is known to use this difference of received frequencies for separating the signals from the echoes of the reverberation. 
     Thus, in the sonar-type electro-acoustic apparatuses, a band rejection filter is used in order to attenuate the reverberation. This filter generally has a fixed band width and a central frequency that varies as a function of the frequency shift caused by the Doppler effect due to carrier movement, i.e. as a function of the speed of this carrier. A filtering device of this type, applied to a torpedo sonar, is described in U.S. Pat. No. 3,723,954. 
     In this device the bandwidth of the rejection filter is calculated as a function of an average speed. Then, this band width varies with speed, the observation direction, i.e. the direction of the path and, as will be seen, with the width of the directivity lobe of this path. 
     SUMMARY OF THE INVENTION 
     The filtering device and method according to the present invention has the advantage, with respect to the prior art, of optimizing the reverberation rejection filter as a function of the speed of the carrier, the directive properties of the sonar, and the direction of the target. 
     For a torpedo sonar, the average level of the secondary lobes remains high and the elevation inclination varies at any moment. This means that the level of reverberation received is due mainly to the border-line reverberations: sea bottom and, above all, surface. 
     According to the invention a filtering device for each signal path calculates, at different instants of signal reception, the extreme frequencies f max  and f min  in order to better eliminate the reverberation signal due to the surface and/or the sea-bed. To accomplish this, angles of bearing and elevation are caused to intervene in the calculation f max  and f min . 
     Two embodiments are described, with a certain number of variants: 
     a temporal analogical or digital filtering of each signal path; 
     a filtering by elimination of spectral-lines, after special analysis of each path signal. 
     This latter embodiment is advantageous in the case where the processing of the signal paths consists in determining an acoustic direction for the search or follow-up of a target, without needing to display the signals: this is the case in torpedoing. 
     A useful simplified variant is also described wherein the elevation angle of the carrier does not vary, i.e. when the filtering device is applied to a sonar of a hull or a sub-marine. 
     According to the invention, a filtering device adapted for signals received by an active sonar mounted on a carrier vehicle travelling at a speed V measured by a speed collector, includes signal path formation circuits B i  corresponding to distinct directions, control means for filtering each signal path, and calculating means to receive a speed signal V and control the filtering means in order to eliminate in the signal path signals the Doppler frequencies due to the speed of the carrier. The calculating means also receive signals S i  emitted by the path formation circuits in order to designate the filtered path, and the said calculating means determines the Doppler frequencies to be eliminated by taking into consideration furthermore the orientation and the width of the filtered path. 
     Other particularities and advantages of the invention will become evident in the following description given by a way of non-limitative example and elaborated with respect to the annexed figures which represent: 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIGS. 1 and 2 the underwater space insonified by a torpedo; 
     FIG. 3 the diagram of a sonar and its filter device according to the invention; 
     FIG. 4 the diagram of a variant of the invention of the sonar of FIG. 3; 
     FIG. 5 the graph of the spectrum of path signals, obtained at the output of device 53 of FIG. 4; 
     FIG. 6 the envelope of the spectrum of a path signal; 
     FIG. 7 the action of a comb filter on the signal of FIG. 6; 
     FIG. 8 diagram of a simplified sonar according to FIG. 3; 
     FIG. 9 diagram of an embodiment of a sonar and its adaptable filtering device using a microprocessor. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIGS. 1 and 2 represent a torpedo 10 maneuvering underwater and the insonified volume 11 corresponding to an instant t of reception for an emitted pulse of duration T. At this instant t of reception, volume 11 corresponds to a distance r equal to c(t-t o )/2, t o  being the emission instant and c the speed of sound. The insonification of the surface and the seabed occurs in two areas 12 and 13 which have the shape of crowns around the vertical axis zz&#39; and have a width proportional to cT/2. These areas define the reverberating zones. 
     In taking, as example, the case of the surface reverberation, zone 12 has an average elevation angle S R  that is equal to arc sin H/r (equation (1) in which H is the depth of the torpedo. The angular width in elevation across the surface reverberation zone 12 is δS. 
     The spectrum of the parasitic signal received in a given sonar path at instant t depends on: 
     the distance r; 
     the duration of pulse T and its spectrum; 
     the emission and reception directivities of the path in azimuth between 0 and 2π; 
     the emission and reception directivities of the path in elevation for S between S R  +δS/2 and S R  -δS/2. 
     inclination α of the carrier with the horizontal; 
     the reverberation index of each elementary part of the reverberation zone (the index depends, for the surface, on the state of the sea). 
     The knowledge of this spectrum at each instant allows a calculation of, for a given path signal, the maximum and minimum frequencies between which the level exceeds the given value. These frequencies constitute cut-off frequencies f min  and f max  of the rejection filter, allowing it to better eliminate, at this instant and for this path, the disturbing signal due to the surface reverberation. 
     According to the same principle, by considering the reverberation intercepted on the sea-bed and by replacing the depth H by the distance to the bed Z, the extreme frequencies of the rejection filter can be determined, allowing better elimination of the disturbing signal due to the reverberation of the sea bed. 
     In a first embodiment, these limits f min  and f max  are calculated in an approximate way. 
     In the first approximation, it is considered that the width of the reverberating zone in bearing 1 is limited to the angular width of the principal lobe δθ i  of the product of the emission-reception directivities in azimuth, width taken, for example, at 3dB attenuation of the maximum. Always in the first approximation only the directions in elevation comprised in the principal lobe of the product of the emission-reception directivities in elevation, width taken also, for example, at 3dB attenuation of the maximum, are considered. For the other directions, it is considered that the reverberation level received is slightly high and that elimination has not been performed. 
     In referring to FIGS. 1 and 2, at a given instant the reverberation level received comes from surface diffusion (for this example) the directions of which are comprised in an angular section delimited by δ.sub.θi and δS. The direction corresponding to r is defined by bearing θ i  of the path considered with respect to axis xx&#39; of the torpedo and by elevation S R  -a in which α is the inclination of axis xx&#39; with respect to the horizontal. Since, with this approximation, it is considered that the reverberated signal is produced by diffusers, the directions of which are comprised in bearing between θ i  -αθi/2 and θ i  δ0i/2, and in elevation S R  -α-δS/2 and S R  -α+δS/2, the expressions f max  and f min  can be determined without having to intervene in the reverberation spectrum of the signal. 
     If v is the speed of the carrier, f 0  the emission frequency and c the speed of sound in water, the shift of frequency Δf, due to the Doppler effect, is given, in the direction of the movement of this carrier, by: 
     
         Δf=2v/cf.sub.0 =K                                    (2) 
    
     In any direction defined by angles θ and S, this shift will be given by: 
     
         Δf=K cos Sωθ                             (3) 
    
     Consequently, the expressions giving f m/n  and f max  will be: ##EQU1## in which S R  is given by (i) and δS by: 
     
         δS=tg S.sub.R ×cT/2r                           (5) 
    
     An embodiment using these expressions (4) is represented in FIG. 3. 
     An antenna 40 emits pulses at frequency f 0  and receives pulses at frequency f R  shifted from f 0  by the Doppler effect with f R  =f 0  +Δf. In a manner known per se, signals S A  supplied by the reception antenna are applied to path formation circuits 41 in which these signals are delayed and added up in order to form angular paths in several directions. 
     Each path signal, such as B, is demodulated in a synchronous way around a frequency f R  =f 0  +Δf by multiplication by two signals in quadrature cos (2πf R  t) and sin (2πf R  t). Frequency f R  is supplied to two multipliers 421 and 422 by a local oscillator 43, which is, for example, a voltage control oscillator (VCO in Anglo-American literature). This oscillator 43 receives value Δf to control the frequency variation. The two components of the demodulated signal are filtered in two identical high-pass filters 441 and 442, the cut-off frequency f c  of which is, according to the invention, a function of the direction and the width of the path. 
     In taking as parameters: ##EQU2## the following relationships are obtained from equations (4): ##EQU3## 
     The values of Δf and f c  are supplied by an assembly of calculation circuits 45 that receive: 
     vehicle speed v, distance to the sea-bed Z, depth H and trim α supplied by suitable measurement devices or pick-ups 410, 411, 412, 413 installed on the torpedo; 
     a signal HO delivered by a clock 415, in order to obtain the measurement of time t that follows from emission instant t o  of each sonar pulse; 
     the values of the other parameters necessary for the calculation of Δf and f c  : the duration of pulse T, the emission frequency f 0  and the speed of sound in water c. 
     Each path signal Bi is indicated by a digital signal S i  that supplies the values of θ i  and δθ/2 contained in memory 450. 
     Circuit 451 calculates from signals HO, t 0  and c, the value of r according to formula r=c(t-t 0 )/2. 
     Two identical circuits 452 and 453 calculate, from this value or r and signals Z and H two values of angle S R , corresponding respectively to the reflection on the sea-bed and the surface, according to equation (1). Indeed, it is only possible to choose between the reflection on the surface and that on the sea-bed after having made the two calculations. 
     Two identical circuits 454 and 455 calculate, for the values of SR, and from the values of r, T and c, the two corresponding values of δS, according to formula (5). 
     Two subtractors 456 and 457 deduct from the two values of S R  the value of angle α. 
     Two adders 458 and 459 add for the two values of S R , the two corresponding values of δS to the values of SR-α obtained at the output of subtractors 456 and 457. 
     Furthermore, two subtractors 460 and 461 substract values δS from values of S R  -α. 
     The value of δθ i  /2 read in memory 450 is respectively added from an adder 462, and substracted in a subtractor 463, to the value of θ i  read from this same memory. 
     Six circuits 464 to 469 calculate the cosines of angles obtained at the outputs of circuits 459 to 463. 
     The cosines of the elevation angles are multiplied with cosines of the bearing angles in four multipliers 470 to 473, to obtain the values of parameters P 1  and P 2  corresponding to depth H and to the distance to sea-bed Z, by applying expression (6). 
     These values are compared with one another in a comparator circuit 474 so as to choose the values of P 1  and P 2  that gives the best rejection: 
     if P 1  (Z)&gt;P 1  (H) and P 2  (Z)&lt;P 2  (H), the circuit supplies P 1  (H) and P 2  (H), and reciprocally; 
     if P 1  (Z)&lt;P 1  (H) and P 2  (Z)&lt;P 2  (H), the circuit supplies P 1  (Z) and P 2  (H). 
     An adder 475 and a subtractor 476 obtain, from these selected values of P 1  and P 2 , (P 1  +P 2 ) and (P 1  -P 2 ). 
     The value of parameter K/2 is obtained, by applying formula (2), from a circuit 477 that receives the values of V, f 0  and c. 
     Two multipliers 478 and 478 thus calculate from K/2, from (P 1  +P 2 ) and from P 1  -P 2 ), Δf and f c  by applying formula (7). 
     In a variant of the invention represented in FIG. 4, the fact is used that the filtering of a signal is shown by the elimination of spectral lines corresponding to the cut band. Thus, a spectral analysis is carried out on each path and values f max  and f min  are converted into numbers of spectral lines that surround the part of the spectrum to be eliminated. 
     Paths B i  and the digital signal that indicates them are obtained as previously described. 
     Thereafter a change of frequency of each path signal is carried out by pulsation in a multiplier 50 with a local frequency OL. This technique lowers by lowering the carrier frequency, the sampling frequency F ech  necessary and thus diminishes the number of special analysis points that will be carried out according to the technique known as FFT. After analog-digital conversion in a converter 51 at rate F ech , the path signals are stocked in a memory 52 so as to be successively treated by spectral analysis, path after path. The spectral analysis is carried out in a FFT analyser 53 over a period equal or close to pulse duration T. 
     Calculation circuits 5 are identical to those of FIG. 3, except that at the output of comparator 474, the two values of P 1  and P 2  are multiplied by K and added to f 0  in order to obtain the values of f min  and f max . 
     Spectrum analyser 53 supplies sequentially provides instantaneous spectrums of all the paths B 1  to B N  as diagramatically represented in FIG. 5. The analysis band corresponds to the maximum Doppler shifts of the goals that are estimated a priori. 
     Circuits 45 supply values f min  and f max . These values allow one to obtain, by reading in a table contained in a memory 55, the numbers n 1  and n 2  of the extreme spectral lines between which the lines must not be taken into account. These numbers n, and n 2  are supplied to detection devices 4 that detects if the remaining spectral lines exceed a certain threshold. 
     Thus, as represented in FIG. 5, the spectral lines between n 1  and n 2  are eliminated, thereby removing a false detection since these lines exceed the threshold S. On the contrary, a target is indeed in paths B 1  and B 2  but not in path B N . 
     In a more elaborate embodiment, the level of all the spectral lines is taken into account. 
     Given a diffusion direction θ,S with respect to the direction of carrier XX&#39;, frequency f i  of the reverberated signal received by the carrier is f i  =f o  +K cos θ cos S. Corresponding to this frequency is a spectral level N i  that can be calculated for each path as a function of the following parameters: 
     instant t j , or distance r, 
     trim α, 
     depth H, 
     distance to the sea-bed Z. 
     To do this, consideration must be taken of the attenuations due to the functions of directivity in bearing and in elevation, at the emission and reception, of the track V i  involved. 
     To each value of (θ,S) corresponds a spectral level N i  and the spectrum is obtained by calculating the levels N i  for the whole of the directions. θ,S correspond to the reverberating zone, i.e. such that 0≦θ&lt;2π and S r  -δS/2 ≦S≦S R  +δS/2. 
     Surfaces 12 and 13 are decomposed into elementary parts in order to supply the calculation pace in θ and S. Between the spectrums taken at different instants, the attenuation corresponding to distance r is introduced. 
     A collection of spectra is obtained such as, for example, that whose envelope is indicated in FIG. 6. The extreme frequencies for a speed v of the carrier are f i  =f 0  -K and f 2  =f 0  +K. 
     The intensity of the reverberation signal is proportional to f 1  ∫ 2  N i  df. By rejecting indiscriminately all the band of f 1  to f 2  there are risks of masking the target if this target corresponds to a weak Doppler effect. According to the invention, is thus established in the form of a spectrum that does not take into account the levels above a given value N S . For this, a battery of numbered connected pass-band filters is considered having a constant width Df, as represented in FIG. 7, and the numbers of the filters in which the level is below N S  are determined, for example filters 1, 4, 5, 16. This operation is carried out for all the collection of spectra, calculated as indicated herein-above. For each path V i , a series of numbers of filters is thus disposed in function of v, H, Z, α and t. 
     Spectral analysis FFT used in the variant of the first embodiment, is equivalent at each instant to the action of a battery of associated pass-band filters having a constant width Df substantially equal to the inverse of the duration of the analyzed signal. 
     The diagram of this second embodiment is that of FIG. 4 except that calculator 45 and circuit 55 are replaced by an addressable memory containing a value table. This table contains numbers of spectral lines to be eliminated, that are previously determined by calculation of each line and for different values of measurement parameters v, H, Z, α and t j , and from the directivity functions involved between 0 and 2π for bearing θ and between -π/2 and π/2 for elevation S. 
     According to the number of path S i  and the values of the measurement parameters, the table is addressed in order to run out the lines to be eliminated in the spectra that are supplied to detection device 54. 
     A simplified variant of the invention, described in connection with the first embodiment, consists in not taking into account elevation angle S by making cos S=1. 
     This condition supposes that angle S, between the diffusers involved and the direction of the carrier, remains small as to the absolute value, and with respect to the bearing angle since the sine only varies rapidly for stable angle values. 
     In order to meet this condition, it is necessary: 
     that the elevation angle of each path is small; 
     that the angular width in elevation of each path is small; 
     that the reverberating volume is strictly limited to the principal directivity lobe of each path; 
     that the elevation angle of the carrier does not vary. 
     Practically, these conditions are more or less satisfied when the filtering device is applied to a hull sonar of a surface or submarine ship. 
     In these conditions, the central frequency in direction θ i , and band w f  corresponding to the angular width δθ i , are given by: 
     
         f=f.sub.0 +K cos θ.sub.i                             (8) 
    
     
         δf=2K sin (δθ.sub.i /2) sin θ.sub.i (9) 
    
     The diagram of this variant, represented in FIG. 8, is similar to that of FIG. 3 in which calculation circuits 45 have been simplified. 
     This signal S i  supplies the addresses of values of θ i  and δθ/2 in a memory 450. The value of speed v is supplied by a measurement device 410 and applied to a calculator circuit 44, that receives, furthermore, the emitted frequency value f 0  and the speed c of acoustic waves in water. Circuit 44 supplies at its output the value of K, according to relation (2). 
     The values of θ i  and δθ i  /2 read from memory 450 are applied to calculation circuits 401, 402 and 403, that give at their outputs, respectively, the values cos θ i , sin θ i  and sin (δθ i  /2). A multiplier circuit M i , receiving the values of cos θ i  and of K, supplies the values Δf applied to a controllableoscillator 43. A second multiplier circuit, receiving the values of sin θ i  and sin δθ i  /2, supplies the product of these two values. This product is applied to a third multiplier circuit M 3  that receives the value of K and supplies the value f c  =δf/2 according to relation (9). This value f c  is applied to the controllable high-pass filters 441 and 442 fixing the cutout value at value f c . It is these filters that receive the signals demodulated by multipliers 421 and 422 and filtered signals X i  and Y i  called complex components are applied to sonar exploitation device 48. 
     Assembly 45 can be a digital calculator of the microprocessor type. 
     FIG. 9 shows an embodiment of a filtering device comprising a microprocessor 500. For this embodiment, N path signals B 1 , B 2 , . . . B i  . . . B N  are applied to a multiplexer 50, that supplies the multiplexed and numerized signal S M  at the rhythm of a clock signal H. Multiplexed signal S M  is treated by microprocessor 500, that also receives the clock signal H and the speed signal v of the carrier. From this value v, the microprocessor calculates for each path the central frequency f R  and cut-off frequency f c . The microprocessor sorts the samples of signal S M  then executes the synchronized demodulation and filtering by a digital filtering program. 
     Furthermore, microprocessor 500 supplies the complex components of paths such as X i  and Y i  applied to the sonar output device. 
     Such an utilization of a calculator in order to carry out the different filtering functions of the sonar can be applied to other embodiments and their variants.