Output control circuit having continuously variable drive current
BACKGROUND OF THE INVENTION 
     The present invention relates to the sizing of transistors in integrated circuits to optimize the efficiency thereof over a broader range of operating parameters. The preferred embodiment of this invention relates to an integrated circuit memory device, specifically a dynamic access memory (&#34;DRAM&#34;), and particularly the output circuit thereof. 
     Integrated circuit memory devices, (sometimes called &#34;chips&#34;) store data and provide it to an output of the chip. The memory can be a static or dynamic RAM, or other types of memory such as read-only memory (&#34;ROM&#34;). Generally speaking, the prior art output circuits for such memories usually include two large field effect transistors (&#34;FETs&#34;) at the output, and data that has been obtained from the memory cells within the chip are applied to these output transistors to drive a fairly substantial current at the output node. The capacitance associated with the output node is typically on the order of 50 picofarads. In the past, the time to drive a data output node was a manageable percentage of the total access time of a DRAM. With increasing memory capacity, however, and decreasing access times, it is important to keep the necessary time to drive the output node relatively small. Thus, it is important to drive the output pin or output node with a relatively large current provided by the output stage of the chip. 
     The design of any integrated circuit must necessarily take into consideration the operating parameters and restrictions imposed on the entire chip. A chip or integrated circuit will be connected to receive a source of operating voltage which in the field of MOS transistors is generally called V CC . The operating voltage V CC  may be 5.0 volts (nominally), but there is no certainty that this power supply voltage will stay constant. It must be assumed that there will be variations in V CC , and integrated circuits in the DRAM field at least must meet specifications for the power supply voltage going higher than nominal. It may safely be assumed that the design must accommodate a power supply voltage of as low as 4.5 volts or as much as 5.5 volts (or more). These are referred to as &#34;low&#34; and &#34;high&#34; voltages, with the &#34;nominal&#34; voltage between them. 
     Whenever the power supply voltage rises to high voltage the transistors inherently work faster and draw more current. The same result occurs when temperature declines--and a wide temperature range is another operational requirement that must be considered in the design of the chip. The normal operating temperature can be referred to as &#34;nominal,&#34; between low and high extremes. In order to accommodate the high currents which result at the extremes of the operating ranges specified for the chip, transistors are constructed to draw known currents at those extreme parameters. The result has been that the circuitry provides somewhat less than optimum performance at operating conditions that are less extreme (or nominal). With regard to the output circuit, in the past where the output circuitry is sized to draw maximum current at six volts, for example, the circuit would draw less current at four volts. By sizing the transistors in this prior art fashion, the current delivered when the power supply voltage is four volts is less than the current delivered when the power supply voltage is six volts. Accordingly, it has taken a longer time to drive the data output node with the power supply at four volts than at six volts. 
     It will be seen therefore that this aspect of integrated circuit design militates against achieving maximum performance at the nominal power supply output voltage. This factor unduly limits the speed of DRAMs since significant time is spent driving the data output node in the output circuit by using prior art techniques. 
     Accordingly, the main object of the present invention is to provide data output circuitry which can operate at high efficiency in nominal operating voltage specifications and temperature specifications even though it is designed to accommodate higher power supply voltages and lower temperatures. As used herein, &#34;nominal&#34; voltage is not limited to 5.0 volts, but for the expected, normal, typical voltage of the power supply. 
     It may generally be stated that an object of the present invention is to provide an improved output control circuit for an integrated circuit memory. 
     A more specific object of the present invention is to provide an output circuit so that the output transistors can be sized for maximum speed at a less extreme power supply voltage than the circuit is required to accommodate. Illustratively, if the power supply voltage is nominally five volts but may rise to six volts, an object of the present invention is to provide maximum speed at five volts instead of maximum speed at six volts. 
     SUMMARY OF THE INVENTION 
     The foregoing and other objects are achieved by providing a circuit so that the current drawn by the output transistors of an output control circuit in an integrated circuit memory is lower at the higher power supply voltage than it is at the nominal power supply voltage. This is achieved preferably by developing a voltage reference which is relatively independent of the power supply voltage V cc  and by combining it with V cc  to provide a further voltage which is applied to the control electrodes of output transistors. Preferably the output transistors are sized for maximum current at the minimum power supply voltage. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In describing the present invention, reference is made to the accompanying drawings wherein: 
     FIG. 1 represents an output control circuit according to the present invention; 
     FIG. 2 is a schematic diagram showing how the input V CC1  is derived for use in FIG. 1; 
     FIG. 3 is an illustration showing the relationship between a voltage V 1  to V CC  in FIG. 2. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 1 illustrates an output control circuit according to several aspects of the present invention. Output control circuit 10 receives an input from a further circuit 12 which is illustrated in FIG. 2. FIG. 3 illustrates the relationship between the voltage at a node in FIG. 2 and the power supply voltage V CC . 
     The circuit of FIG. 1 receives as inputs, to data receiving circuit complementary data D and D BAR, and it provides a data output D out . The only other essential inputs to circuit 10 are the power supply voltage (operating voltage) V CC , a reference potential (ground), and a voltage marked V CC1 . The latter voltage is derived from V CC  by the circuit 12 shown in FIG. 2. 
     FIG. 2 receives a voltage reference input V REF  at an input node 14, which is coupled to the gate electrode of a p-channel transistor 16. The source of transistor 16 is coupled to the power supply voltage V CC  at node 18. The drain of transistor 16 is coupled to a node 20. The reference voltage is also applied to the gate electrode of a high impedance transistor 22. Transistor 22 is an n-channel device having its drain coupled to node 20 and its source coupled to ground. Node 20 is coupled also to the drain of a further n-channel FET 24, the gate electrode and source of which are both grounded. Node 20 provides a voltage V 1  which is applied to the gate electrode of a p-channel transistor 23. The drain 26 of transistor 23 provides the output V CC1  used in FIG. 1. The source of transistor 23 is a node 28 which is coupled to the power supply voltage. 
     It will be understood that the V REF  input to node 14 of FIG. 2 is a reference voltage that is substantially independent of V CC . A circuit for generating such a voltage is shown and described in U.S. patent application Ser. No. 07/644,904 filed herewith, entitled REFERENCE GENERATOR FOR AN INTEGRATED CIRCUIT, now U.S. Pat. No. 5,117,177, issued May 26, 1992. 
     In FIG. 2, transistor 16 is sized relative to transistor 24 so that voltage at node 20 therebetween rises faster than rises occur in V CC . FIG. 3 illustrates the relationship between V CC  and V 1 . It will be recalled that the conductivity of a p-channel transistor is dependent on the gate-to-source voltage. Specifically, if the gate voltage is below the source voltage by at least one threshold voltage T T , then the transistor is ON (conductive). Consequently, considering transistor 23 of FIG. 2, if V 1  is six volts when V CC  is six volts, then transistor 23 is OFF and no current passes therethrough. If V 1  is zero volts when V CC  is four volts, then current will be drawn through transistor 23 from V CC  to node 26. If V 1  is four volts when V CC  is six volts, current is still drawn through transistor 23. The impedance of transistor 23, however, is greater at six volts than at four volts because V 1  has risen faster than V CC  has risen (where V CC  rose two volts, V.sub. 1 rose four volts.) Since V 1  rises faster than V CC , it allows current through transistor 23 to be higher at four volts than at six volts. 
     Thus it will be appreciated that the circuit of FIG. 2 outputs a derived voltage or a modified power supply voltage which, over a range of power supply voltage values that can be supplied to or encountered in the device, rises faster than the power supply voltage rises. Preferably, the relationship between the modified power supply voltage and the power supply voltage is quadratic but could be linear so long as the derived power supply voltage is a function of the power supply voltage and increases faster than the power supply voltage increases. 
     The circuit of FIG. 1 combines the derived power supply voltage V cc1  with the power supply voltage V cc  at an output transistor so that maximum current can be drawn by the output transistor at four volts rather than six volts, illustratively. Referring to FIG. 1, the derived power supply voltage is V cc1  is applied to an input 30. The complementary data input (DATA BAR) 32 together with the derived power supply voltage are applied to an inverter 34. The inverter typically comprises a CMOS pair of FETs with their source-drain paths coupled between ground and a source of operating voltage applied here at input 30. The input 32 is coupled to the gate electrodes, and the output node of the inverter is coupled to the drains of the FETs which operate in push-pull fashion. The data input is applied to an input 36 of a further inverter 38 which also receives the derived power supply voltage V cc1 . The output of inverter 38 is node 40. 
     As shown in FIG. 1, a set of transistors is coupled to node 40. Specifically, node 40 is coupled to the gate electrode of an n-channel transistor 42. From there it is further coupled via a resistor 44 to the gate electrode of another n-channel transistor 46. Additionally, via a resistor 48, that gate electrode is further coupled to the gate electrode of another n-channel transistor 50. Yet a further resistor 52 couples that gate electrode to the gate electrode of a further n-channel transistor 54. The drain electrodes of transistors 42, 46, 50 and 54 are all coupled to a node 56. The source electrodes of transistors 42, 46, 50 and 54 are all grounded. 
     This set or network of transistors and resistors provide RC coupling in a different fashion than used in the prior art. The capacitance referred to herein is the capacitance of the gate electrodes of the transistors. Previously, in order to decrease the peak current, an output circuit used an output transistor having a long distributed resistance gate, illustratively 500 microns long. The present configuration has been found to be more efficient than simply applying a data signal to a pull-down transistor with a long resistance gate. This has not been found to be true for the pull-up transistor 60 which is also shown in FIG. 1. Transistor 60 is preferably an n-channel transistor having its drain coupled to the power supply voltage V cc  and its source coupled to node 56, which provides the output D out . Transistor 60 is responsively coupled to the inverter 34 by virtue of a node 62 coupled to the output of inverter 34 and to the gate electrode of transistor 60. 
     If the circuit of FIG. 1 were drawn according to the prior art, it would not use the derived power supply voltage V cc1 , nor would it use the discrete RC network. Pull-up transistor 60 would be sized to draw its maximum current at the highest specified power supply voltage, illustratively six volts. However, since the nominal power supply voltage is below the highest permissible power supply voltage, the speed of transistor 60 would be slower at the nominal power supply voltage than at the high power supply voltage. It would take a longer time at the nominal power supply voltage to drive the capacitance coupled to output node 56 to the proper data value than it would at the high power supply voltage 
     By virtue of the present invention, however, transistor 60 does not need to be sized for maximum current at the high power supply voltage. Instead, as described above, the derived power supply voltage V cc1  rises faster than V cc  and the current through transistor 23 at the power supply voltage is greater than the current therethrough at the high power supply voltage. 
     The operation of transistor 23 provides a corresponding operation in the circuit of FIG. 1. Specifically, the rise time of node 62 is slower at six volts (high) than it is at four volts (low). This is because less current is drawn at six volts than at four volts. Accordingly, more current is drawn through output pull-up transistor 60 at the low power supply voltage of four volts than at the high power supply voltage which is illustratively six volts. 
     Consequently, transistor 60 is sized for maximum speed at the low voltage of four volts, thereby improving the operation of the output circuitry for an integrated circuit memory device. Some of the sizes of the devices in the drawings are as follows, giving transistor width/lengths in microns: 
     16=8/3 
     22=2/20 
     23=250/1.5 
     24=3/3 
     42=72/1.1 
     44, 48, 52=1600 ohms 
     46=36/1.1 
     50=55/1.1 
     54=210/1.1 
     66=50 pf. 
     V REF  =3.5 V