TTL to CMOS translating input buffer circuit with dual thresholds for high dynamic current and low static current
TECHNICAL FIELD 
     This invention relates to a new input buffer circuit having an input for receiving data signals at TTL logic high and low potential levels and an output for delivering data signals at CMOS logic high and low potential levels. The invention provides an expanded first stage with enhanced dynamic pullup current during LH transitions at an intermediate output node in response to the HL transitions at the input to increase switching speed. The invention also reduces static current I CCT  by restricting pullup current through the first stage during a steady state high potential level data signal H at the input and a corresponding steady state low potential level data signal L at the intermediate output node. The expanded input stage switches data signal voltage levels at a lower TTL first threshold voltage level and switches the enhancing pullup current at a higher specified second threshold voltage level. In the expanded first stage, pullup current is controlled from the input by a network of multiple pullup transistors coupled in parallel and in series. The dual threshold input buffer circuit parameters are scaleable and programmable according to the circuit application. 
     BACKGROUND ART 
     A prior art TTL to CMOS translating input buffer circuit consisting of first and second CMOS inverter stages is illustrated in FIG. 1. The first inverter stage P1,N1 is coupled between the input V IN  and an intermediate output node m1. The second inverter stage P2, N2 is coupled between the intermediate node m1 and the output V OUT . The inverter stages P1,N1 and P2,N2 are coupled between high and low potential power rails V CCQ  and GNDQ. The high potential power rail V CCQ  voltage at, for example 5.0 v, and the low potential power rail GNDQ voltage at, for example 0 v, represent the CMOS logic high and low potential levels. 
     TTL high and low potential level data signals, typically 2.0-2.4 v high and 0.4-0.8 v low are applied at the input V IN . The ratio of respective channel widths of the PMOS pullup transistor P1 and NMOS pulldown transistor N1 of the first inverter stage is skewed to provide a TTL switching threshold voltage level at the input V IN  of typically 1.5 v. To achieve this TTL switching threshold voltage, the ratio of channel widths P1/N1 is typically 1/4. 
     An LH transition to a TTL high potential level data signal H at the input V IN  turns on the NMOS pulldown transistor N1 to cause a low potential data signal L at the intermediate output node m1. The TTL high potential signal H is not sufficient however to complete turn off of PMOS pullup transistor P1 causing an unwanted static current or crowbar current I CCT  to flow through P1 and N1 during a steady state low potential data signal L at the intermediate node m1. Transistor P1 is therefore sized for a small channel width to restrict and limit the undesirable power dissipating static current I CCT  to an acceptable specified level. 
     Typical values for channel widths of P1/N1 are, for example, 25 μ/100 μ for the same channel length. While this skewed ratio and small size P1 channel width limits static current I CCT  to a specified level, it slows the LH transition from low to high potential level at intermediate output node m1. Data signals reach the CMOS logic power rail voltage levels of 0 v and 5.0 v at the intermediate node m1 but with reduced speed and with unwanted power dissipation. 
     Intermediate node m1 drives the second CMOS inverter stage P2,N2 which is selected to have channel widths in a standard ratio range of for example 1/1 to 1/2. Example values for channel widths of P2,N2 are, for example 150 μ/150 μ for the same channel length. The second CMOS inverter stage P2,N2 with standard ratio channel widths switches at the CMOS threshold voltage of e.g. 2.5 v and completes the non-skewed translation to CMOS logic high and low potential level data signals at the output V OUT . Output signals at V OUT  are available to drive other CMOS or BICMOS circuits. 
     OBJECTS OF THE INVENTION 
     It is therefore an object of the present invention to overcome the pullup current path restriction of a reduced channel width first inverter stage pullup transistor in TTL to CMOS translating input buffer circuits. A purpose of the invention is to provide enhanced dynamic pullup current during LH transitions at the intermediate output node in response to HL transitions at the input to increase switching speed. 
     At the same time, another object of the invention is to restrict the pullup current through the first inverter stage during steady state low potential data signals L at the intermediate output node to meet reduced static current I CCT  specifications. 
     DISCLOSURE OF THE INVENTION 
     In order to accomplish these results, the invention provides a new input buffer circuit with an expanded first stage or input stage. The first stage is constructed to provide dual input switching voltage thresholds at the input. The first stage output pullup and pulldown circuit switches at a relatively lower first threshold voltage level. A pullup enhancer circuit switches at a relatively higher second threshold voltage level. 
     According to the preferred example embodiment of the invention, the first stage output pullup and pulldown circuit of the expanded first stage is constructed for switching dynamic current at an output node m1 at the relatively lower first threshold voltage level for data signal transitions between high and low potential levels at the output node. The pullup enhancer circuit is constructed for switching static current I CCT  through the intermediate output node m1 at the relatively higher second threshold voltage level to reduce static current I CCT  during a static low potential level data signal at the output node m1. The relatively lower first threshold voltage level is selected to be substantially at a TTL input switching threshold while the relatively higher second threshold voltage level is selected to be substantially at a CMOS input switching threshold. 
     Broadly conceived, the invention thus provides a new method of buffering input data signals at the input of an input buffer circuit having a first stage output pullup and pulldown circuit coupled to an output node. The method steps include switching the first stage output pullup and pulldown circuit at a relatively lower first threshold voltage level, and switching a pullup enhancer circuit at a relatively higher second threshold voltage level. 
     The preferred method contemplates switching dynamic current at the output node at the first threshold voltage level for data signal transitions at the output node and switching static current at the output node at the second threshold voltage level to reduce static current I CCT . The first stage output pullup and pulldown circuit may be constructed for switching at a TTL input switching first threshold voltage level and the pullup enhancer circuit may be constructed for switching at a CMOS input switching second threshold voltage level. 
     In the preferred example, the invention provides a TTL to CMOS translating input buffer circuit with a first stage expanded pullup circuit Pl including a relatively small channel width static current restricting PMOS first pullup transistor and a relatively large current conducting dynamic current enhancing second pullup transistor. The first and second pullup transistors have primary current paths coupled in parallel to a high potential power rail. A pullup current summing PMOS third pullup transistor is coupled to the intermediate output node in series with the parallel coupled first and second pullup transistors. The first, second and third pullup transistors are coupled in the input buffer circuit for operations substantially in phase. 
     According to the invention the first and third pullup transistors have control gate nodes coupled to the input and form part of the first stage output pullup and pulldown circuit. The invention also provides a dynamic current enhancing and static current reducing control circuit coupled between the input and a control gate node of the current enhancing second pullup transistor. The control circuit is constructed for turning on the second pullup transistor to deliver enhanced sourcing current during an LH transition at the intermediate output node m1 to increase switching speed. The control circuit also turns off the second pullup transistor during a static low potential level data signal L at the intermediate output node m1 to reduce static current I CCT . 
     In the preferred example the dynamic current enhancing and static current reducing control circuit is provided by first and second inverting CMOS stages. The control circuit inverting stages are composed of small channel width complementary PMOS and NMOS transistors for fast operation of the second pullup transistor substantially in phase with the first and second pullup transistors. The channel widths of the complementary PMOS and NMOS transistors of the control circuit inverting stages are formed with PMOS and NMOS channel width ratios selected for switching the current enhancing second pullup transistor for controlling static current at a relatively higher second threshold voltage level at the input. 
     The input stage output pullup and pulldown circuit includes a pulldown circuit which is provided by a relatively large channel width pulldown transistor. The large channel width pulldown transistor is coupled to the input with the first and third pullup transistors for switching dynamic current to effect a transition of the data signal potential level at the intermediate output node m1 at a relatively lower first threshold voltage level. 
     A feature of the dual threshold voltage first stage or input stage of the input buffer circuit of the invention is that the enhancing pullup current through the second pullup transistor is controlled separately from the switching of data signal voltage levels. The channel width or current conducting capacity of the dynamic current enhancing second pullup transistor is substantially greater than the channel width of the static current restricting first pullup transistor. Upon turning off the second pullup transistor, completely, the static current I CCT  through the first pullup transistor is therefore limited to an acceptable specified level. The channel width of the third pullup transistor is greater than that of either the first or second pullup transistor for combining the enhanced and restricted pullup current during LH transitions at the intermediate output node m1. 
     According to the invention the first, second and third pullup transistors of the input buffer circuit first inverter stage are selected to provide an equivalent pullup circuit P1 effective channel width or current carrying capacity for the first stage pullup circuit substantially less than the channel width or current carrying capacity of the first stage pulldown transistor circuit N1. This skewed ratio of effective pullup channel width to pulldown channel width for example in the order of 1/4 provides the desired TTL first switching threshold voltage level at the input for switching dynamic current to effect a transition of data signal voltage levels at for example 1.5 v. At the same time, the channel width ratios of the complementary PMOS and NMOS transistors of the control circuit inverter stages are selected to switch the dynamic enhancing pullup current through the second pullup transistor at a higher specified switching threshold voltage level for example in the range of 2.5 v to 3.4 v to control static current. 
     An advantage of this input inverter stage according to the invention is that the current enhancing second pullup transistor provides substantially increased pullup current for LH transitions at the intermediate output node in response to HL transitions at the input with increased switching speed. At the same time, the static current restricting first pullup transistor limits static current during steady state low potential level signals L at the intermediate output node to an acceptable circuit specification. 
     Another feature of the invention is that the effective channel width ratio for the input stage pullup circuit P1 to pulldown circuit N1 operates at approximately a standard skewed ratio value of 1/4 during dynamic switching of an LH transition at the intermediate output node in response to an HL transition at the input. Furthermore the absolute values of the effective channel widths may be substantially greater, for example, four times greater than the absolute channel dimensions of a conventional input inverter stage, for substantial reduction in propagation delay and increase in switching speed. However during a steady state low potential level data signal L at the intermediate output node, the ratio of effective channel widths of the input stage pullup circuit to pulldown circuit operates in a substantially smaller ratio range of for example 1/8 to 1/16 for suppressing static current I CCT  in order to meet required circuit specifications. 
     An advantage of the invention is that the respective channel dimensions and channel width ratios of the input buffer circuit input stage CMOS transistors may be scaled and selected for programming a wide range of circuit operating parameters according to the circuit application. For example the first and second dual threshold voltage levels may be programmed according to the channel width ratios selected for complementary pairs of PMOS and NMOS transistors and the effective or equivalent channel ratios of the input stage pullup and pulldown circuits P1,N1. Furthermore the absolute value sizes of the channel dimensions may be selected for achieving the desired pullup and pulldown current drive, propagation delay, and switching speed. 
     In an alternative embodiment of the invention, the pullup current summing third pullup transistor function is divided between current summing PMOS third and fourth pullup transistors. The third pullup transistor is coupled in series with the current enhancing second pullup transistor. The fourth pullup transistor is coupled in series with the static current restricting first pullup transistor. The pullup current summing third and fourth pullup transistors are coupled with primary current paths in parallel to the output node m1 of the expanded first stage. An advantage of this circuit arrangement is that it reduces node capacitance between the first pullup transistor and the summing fourth pullup transistor for higher dynamic switching speed. 
     The invention is applicable to both CMOS and combined bipolar and CMOS (BICMOS) circuits. Alternative embodiments are described for both CMOS and BICMOS circuits. 
     Other objects, features and advantages of the invention are apparent in the following specification and accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic circuit diagram of a prior art TTL to CMOS translating input buffer circuit. 
     FIG. 2 is a detailed schematic circuit diagram of a TTL to CMOS translating input buffer circuit according to the invention. 
     FIG. 2A is a simplified circuit diagram of the TTL to CMOS translating input buffer circuit of FIG. 2. 
     FIG. 3 is a graph comparing the propagation delay for the circuits of FIGS. 1 &amp; 2. 
     FIG. 4 is a graph comparing input switching threshold voltage levels at static current I CCT  values for the circuits of FIG. 1 &amp; 2 including comparison with the circuit of FIG. 1 for different channel width values for the pullup circuit PMOS transistor. 
     FIG. 5 is a fragmentary schematic circuit diagram of a BICMOS circuit modification of the input buffer circuit of FIGS. 2 and 2A replacing the PMOS current enhancing second pullup transistor P1E with a bipolar NPN transistor P1E (NPN). 
     FIG. 6 is a fragmentary schematic circuit diagram of another BICMOS circuit modification of the input buffer circuit of FIGS. 2 and 2A replacing the PMOS dynamic current enhancing second pullup transistor P1E with a bipolar PNP transistor P1E (PNP). 
     FIG. 7 is a fragmentary schematic circuit diagram showing an alternative circuit modification of the input buffer circuit of FIGS. 2 and 2A in which the pullup current summing third pullup transistor P1L function is divided between pullup current summing third and fourth pullup transistors P1LA and P1LB coupled in parallel. 
    
    
     DESCRIPTION OF PREFERRED EXAMPLE EMBODIMENTS AND BEST MODE OF THE INVENTION 
     A TTL to CMOS translating input buffer circuit according to the invention is illustrated in FIG. 2, with a simplified equivalent circuit diagram shown in FIG. 2A. The pullup transistor circuit P1 for the input inverter stage P1,N1 has been replaced with a complex network of multiple PMOS pullup transistors P1R,P1E,P1L coupled in parallel and in series. A static current limiting first pullup transistor P1R and a current enhancing second pullup transistor P1E are coupled with primary current paths in parallel to the high potential power rail V CCQ . A pullup current summing third pullup transistor P1L is Coupled to the intermediate output node m1 from the first inverter stage with primary current path in series with the parallel coupled transistors P1R,P1E. 
     The control gate nodes of the static current restricting first pullup transistor P1R and the source current combining third pullup transistor P1L are coupled directly to the TTL data signal voltage level input V IN . Along with pulldown transistor N1L, the pullup transistors P1R and P1L form the first stage output pullup and pulldown circuit. 
     The control gate node of the pullup current enhancing third pullup transistor P1E is coupled to the input V IN  through a control circuit I3,I4 hereafter described. The control circuit turns on the second pullup transistor P1E during an LH transition at intermediate output node m1 and holds transistor P1E on during a steady state high data signal H at node m1. Pullup transistor P1E and control circuit inverter stages I3,I4 form the first stage pullup enhancer circuit. 
     The first pullup transistor P1R has a relatively small channel width comparable to pullup transistor P1 in the prior art circuit of FIG. 1, for example 25 μ, and generates only a restricted sourcing current I SR  in order to limit subsequent static current I CCT . Because the control gate node is coupled directly to the input V IN , it is not pulled up to the CMOS high potential power rail voltage of V CCQ  when a TTL logic high data signal is applied at the input V IN . The partial turn off results in static current I CCT  through the first pullup transistor P1R. 
     The parallel current enhancing pullup transistor P1E is provided with a relatively large channel width, e.g. 175 μ, for generating a relatively large enhancing sourcing current I SE  to increase sourcing current. With a low potential level signal L at the input V IN , P1E is ready to source current. In response to an HL transition at the input V IN , P1E generates the large enhancing sourcing current and speeds up the LH transition at intermediate output node m1. The source current third pullup transistor P1L coupled in series is provided with a sufficiently large channel width, e.g. 200 μ, to accommodate and combine the restricted source current I SE  and enhancing source current I SE  to deliver a large enhanced pullup current I SL  at the intermediate output node m1. 
     The pulldown circuit N1 is provided by a relatively large channel width NMOS pulldown transistor N1L. The transistors of the pullup and pulldown circuits P1,N1 are sized in relation to each other as follows. The equivalent pullup circuit channel width of the network of pullup transistors P1R,P1E,P1L when all the pullup transistors are conducting is selected to provide an input stage ratio of equivalent pullup circuit channel width to the channel width of pulldown transistor N1L of, for example, 1/4. This establishes the desired TTL switching threshold voltage level at the input V IN  of approximately 1.5 volts. In the example summarized in TABLE I, the equivalent pullup circuit channel width of pullup network P1R,P1E,P1L is 100 μ while the channel width of the pulldown transistor N1L is 400 μ. 
     It is apparent that the circuit of FIGS. 2 and 2A permits substantially greater channel width pullup and pulldown circuits P1,N1 than the prior art circuit of FIG. 1. In the example of TABLE I, the current drive capacity of the input inverter stage is 4 times that of the circuit of FIG. 1 for greater output load capacity and higher switching speed. 
     During a steady state low potential data signal L at intermediate output node m1 in response to a TTL high potential data signal at the input V IN , the dynamic current enhancing second pullup transistor P1E is fully turned off as hereafter described, so that unwanted static current is limited to the 25 μ channel width first pullup transistor P1R. Undesirable static current is therefore limited to the same specifications as the circuit of FIG. 1. The effective channel width ratio of the pullup and pulldown circuits P1,N1 for the circuit of FIG. 2 is therefore 1/16 during static current conditions. Operation of the control circuit I3,I4 to achieve this result is described with reference to the example of TABLE I. 
     
                       TABLE I
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Transistor Channel Widths in Microns
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        P1R          25μ
        P1E          175μ
        P1L          200μ
        N1L          400μ
        P3           20μ
        N3           6μ
        P4           6-12μ
        N4           6-12μ
        P2           150μ
        N2           150μ
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     The control circuit at the gate node of PMOS transistor P1E consists of two inverter stages I3,I4. The inverter stage I3 comprises a pair of complementary CMOS transistors P3,N3 and inverter stage I4 comprises complementary CMOS transistors P4,N4. The absolute channel width sizes of the inverter stage CMOS transistors P3,N3, P4,N4 are small for fast switching so that the enhancing second pullup transistor P1E operates substantially in phase with the first and third pullup transistors P1R,P1L which are coupled directly to the input V IN . However, the ratio of respective channel widths of the control circuit complementary CMOS transistors P3/N3 is selected to provide a higher second switching threshold voltage level for switching on and off the second pullup transistor P1E and dynamic enhancing pullup current I SE . In the example of TABLE I the channel width ratio for P3/N3 is skewed to establish a second threshold voltage level at the input of, for example in the range of 2.5 v to 3.4 v. Data signal voltage levels at input node V IN  and output node V OUT  are thus switched at the lower TTL first threshold voltage of approximately 1.5 v while the pullup enhancing current through second pullup transistor P1E is switched at the higher CMOS second threshold voltage of approximately 2.5 to 3.4 v. 
     With a low to high data signal transition LH at the input V IN , the large NMOS pulldown transistor N1L turns on and initiates the high to low data signal transition HL at intermediate output node m1 and the final low to high data signal transition LH at the output V OUT . Switching occurs at the relatively lower TTL first switching threshold voltage level of approximately 1.5 v as illustrated in the graph of FIG. 4. As pulldown transistor N1L turns on there is a jump to a maximum peak in static current I CCT  because the network of pullup transistors P1R,P1E,P1L of the pullup circuit P1 are all still on from the previous low data signal condition L at the input V IN . As the voltage level at the input rises, the first and third pullup transistors P1R,P1L having gate nodes coupled directly to the input V IN  begin to turn off reducing the crowbar current I CCT  as further shown in the graph of FIG. 4. 
     When the data signal voltage level at the input V IN  rises to a relatively higher CMOS second switching threshold voltage level, the control circuit inverter stages I3,I4 turn off the current enhancing second pullup transistor P1E. The PMOS transistor P4 of inverter stage I4 pulls up the gate node of dynamic current enhancing pullup transistor P1E to the CMOS high potential level voltage of the high potential power rail V CCQ  so that it is entirely off. No static current passes through transistor P1E. There is a final drop in the static current I CCT  as shown in the graph of FIG. 4 to the minimum acceptable level of static current passing through the first pullup transistor P1R. The channel width dimension of the static current restricting pullup transistor P1R can therefore be selected to meet the specification for the circuit. 
     The graph of FIG. 4 provides a comparison of the crowbar current I CCT  of the new circuit of FIG. 2 with the prior art circuit of FIG. 1. For purposes of standardized I CCT  measurement and testing, a high potential level signal H of 3.4 v is applied at the input V IN  for the measurement of I CCT  shown in the graph of FIG. 4. Results for the circuit of FIG. 1 are shown with two different channel width dimension values for the input stage pullup transistor P1. It is apparent that the circuit of FIG. 2 combines the advantage of the large dynamic switching current afforded by a relatively large channel width dimension pullup circuit P1 (100 μ) with the advantages of a limited static current comparable to a small channel width dimension pullup circuit P1 (25 μ). An additional small I CCT  component I PRE  which passes through inverter stage I3 is also shown in the graph of FIG. 4. 
     A comparison of the propagation delays for the respective circuits of FIGS. 1 and 2 is shown in the graph of FIG. 3. The circuit of FIG. 2 initiates and completes the data signal switching transitions in reduced time intervals with an increase in switching speed of approximately 10%. This is achieved with the increased dynamic current drive equivalent to a channel dimension ratio of the pullup circuit Pl to pulldown circuit N1 of approximately 100/400, four times greater than the prior art circuit of FIG. 1. Yet static current I CCT  is restricted to the same level as the small channel dimension pullup transistor P1 of FIG. 1. 
     The absolute values of the channel dimensions, and the channel width ratios of the CMOS transistor pairs are scaleable and programmable to achieve whatever dynamic current and static current performance is required for a particular circuit application. An example of an input buffer circuit having an input stage with even greater dynamic current current drive yet with the same restricted static current I CCT  is set forth in TABLE II. 
     
                       TABLE II
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Channel Width Values in Microns
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        P1R           25μ
        P1E           375μ
        P1L           400μ
        N1L           800μ
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     It is noted that in the circuit of FIGS. 2 and 2A, the first stage pullup and pulldown circuits P1,N1, and the second stage P2,N2, are coupled to separate high and low potential power rails V CCQ , V CCP , and GNDQ, GNDP. Such separate power rails are optional and provide noise isolation between input and output stages. The power rails may be entirely separate. Or, alternatively, relative separation of the respective &#34;quiet&#34; (V CCQ ,GNDQ) and &#34;noisy&#34; (V CCP ,GNDP) power rails can be provided, for example, using split lead lead frames as described in U.S. Pat. No. 5,065,224 issued Nov. 12, 1991. 
     The CMOS input buffer circuit of FIGS. 2 and 2A may be modified for BICMOS circuit applications incorporating both CMOS and bipolar transistors. In the example of FIG. 5 the current enhancing second pullup transistor P1E is replaced by a bipolar NPN transistor P1E (NPN). Use of an NPN bipolar transistor P1E (NPN) permits elimination of one of the inverter stages I4 of the control circuit I3,I4 as shown in FIG. 5. All other elements of the input buffer circuit of FIG. 2A remain the same. Because the NPN bipolar transistor P1E (NPN) turns off in response to a low potential level data signal at the base node, it operates in phase with the first and third PMOS pullup transistors P1R,P1L with only a single control circuit inverter stage I3. 
     In the BICMOS circuit modification example of FIG. 6, the PMOS current enhancing second pullup transistor P1E is replaced with a PNP bipolar transistor P1E (PNP). The PNP bipolar transistor is analogous in the logic of its operation with a PMOS transistor and turns on in response to a low potential level data signal at the base node. Both inverter stages of the control circuit I3,I4 are therefore retained for operation of the PNP bipolar transistor P1E (PNP) in phase with the first and third PMOS pullup transistors P1R,P1L. The remaining portions of the circuit of FIG. 6 are the same as the input buffer circuit of FIG. 2A. A further BICMOS circuit modification of the input buffer circuit is that the second stage or output stage CMOS transistors P2,N2 may be replaced with bipolar pullup and pulldown transistors. 
     An alternative embodiment of the input buffer circuit of FIGS. 2 and 2A is illustrated in the fragmentary schematic circuit diagram of FIG. 7. In this example the pullup current summing function of the PMOS third pullup transistor P1L of FIGS. 2 and 2A is divided between current summing third and fourth pullup transistors P1LA and P1LB as shown in FIG. 7. The current summing third pullup transistor P1LA is coupled in series with the current enhancing second pullup transistor P1E. The current summing fourth pullup transistor P1LB is coupled in series with the static current restricting first pullup transistor P1R. The third and fourth pullup transistors P1LA and P1LB are in turn coupled in parallel to the intermediate output node m1. An advantage of the circuit configuration of FIG. 7 is that it reduces capacitance at an intermediate node m2 between the first pullup transistor P1R and the fourth pullup transistor P1LB. This reduced capacitance at node m2 improves dynamic switching speed at intermediate output node 
     While the invention has been described with respect to particular example embodiments it is intended to cover all modifications and equivalents within the scope of the following claims.