Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> The CMOS technology area has quickly grown, calling for a new text--and here it is, covering the analysis and design of CMOS integrated circuits that practicing engineers need to master to succeed. Filled with many examples and chapter-ending problems, the book not only describes the thought process behind each circuit topology, but also considers the rationale behind each modification. The analysis and design techniques focus on CMOS circuits but also apply to other IC technologies. ::: Table of contents ::: ::: 1 Introduction to Analog Design ::: 2 Basic MOS Device Physics ::: 3 Single-Stage Amplifiers ::: 4 Differential Amplifiers ::: 5 Passive and Active Current Mirrors ::: 6 Frequency Response of Amplifiers ::: 7 Noise ::: 8 Feedback ::: 9 Operational Amplifiers ::: 10 Stability and Frequency Compensation ::: 11 Bandgap References ::: 12 Introduction to Switched-Capacitor Circuits ::: 13 Nonlinearity and Mismatch ::: 14 Oscillators ::: 15 Phase-Locked Loops ::: 16 Short-Channel Effects and Device Models ::: 17 CMOS Processing Technology ::: 18 Layout and Packaging <s> BIB001 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> Multi-frequency electrical impedance tomography (MF-EIT) systems require current sources that are accurate over a wide frequency range (1 MHz) and with large load impedance variations. The most commonly employed current source design in EIT systems is the modified Howland circuit (MHC). The MHC requires tight matching of resistors to achieve high output impedance and may suffer from instability over a wide frequency range in an integrated solution. In this paper, we introduce a new integrated current source design in CMOS technology and compare its performance with the MHC. The new integrated design has advantages over the MHC in terms of power consumption and area. The output current and the output impedance of both circuits were determined through simulations and measurements over the frequency range of 10 kHz to 1 MHz. For frequencies up to 1 MHz, the measured maximum variation of the output current for the integrated current source is 0.8% whereas for the MHC the corresponding value is 1.5%. Although the integrated current source has an output impedance greater than 1 MOmega up to 1 MHz in simulations, in practice, the impedance is greater than 160 kOmega up to 1 MHz due to the presence of stray capacitance. <s> BIB002 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> We describe a new method for high-frequency precision sensing. The method combines synchronous detection with chopping in a fully differential architecture that includes an instrumentation amplifier. An integrated circuit implementation of the proposed synchronous chopping demodulator front end was designed and fabricated in a 0.35-mum CMOS process technology and tested with high-frequency inductive position sensors. The measured results show that the new technique offers considerable advantages in terms of offset reduction compared to traditional techniques for these sensors, which rely on a microcontroller to measure the offset before each position measurement is taken. The measured average input-referred offset for the 20 fabricated chip samples is 87 muV at a chopping frequency of 500 kHz when the resonant target is off and synchronous demodulation and transmitter excitation are both applied at 1 MHz. The technique, in addition to improving system resolution and immunity to common-mode interference, allows these high-frequency position sensors to work with multiple targets, thus increasing speed and functionality. <s> BIB003 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> Advances in bioimpedance measurement applications need current drivers with high output impedance in the frequencies ranging from DC to a number of MHz. In particular, it needs greater than 10 MΩ output impedance at low frequencies, e.g., 10 kHz. This paper presents an current driver based on operational transconductance amplifiers (OTAs) using a feedback loop to provide over 10 MΩ output impedance ranging from DC to 10 kHz and 1 MΩ at 1 MHz. In addition, a stable performance over a wide frequency range (<4 MHz) is simulated. The circuit has been designed in a 0.6-μm CMOS process using ±2.5 V power supplies. Simulated results are presented. <s> BIB004 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> Bioelectrical impedance measurements are routinely used in many clinical applications such as lung function monitoring, brain imaging, body segment monitoring and tissue characterization. Advantages of the method include low-cost, simplicity and non-invasiveness. One bioimpedance imaging technique of particular interest is Electrical Impedance Tomography (EIT). EIT works by reconstructing the differences in electrical conductivity inside a body. The design of instrumentation for bioimpedance measurements is very challenging, particularly for applications requiring wideband operation (e.g., imaging of cancer biomarkers). Such applications require wideband current drivers that can deliver accurate and stable sinusoidal current (AC) to the electrode-tissue load. Many current drivers for bioimpedance measurements are based on discrete electronics, mostly using the modified Howland circuit (MHC). However, the accuracy and performance of the MHC depends on the performance of the operational amplifiers used and on the degree of matching of its resistors, making the approach not attractive for integration. In addition, because it uses both positive and negative feedback it can suffer from stability problems and since it employs voltage feedback it suffers from bandwidth limitations. ::: This thesis is concerned with the design implementation and experimental evaluation of novel current drivers for wideband, fully-integrated, bioimpedance instrumentation for EIT applications. Three new current driver circuits are introduced based on the operational transconductance amplifier (OTA) approach, and implemented in sub-micron CMOS technology. The described integrated current drivers overcome many of the limitations of discrete current drivers, i.e., limited precision at high frequencies, stability, and low-bandwidth. Two new open-loop current drivers implemented in 0.35-μm CMOS technology are presented and their performance is analyzed and simulated. One of the current drivers was fabricated as an integrated circuit and its performance was experimentally verified using RC loads, a saline tank electrode phantom and electrodes on the forearm. In order to further improve its performance, a novel negative impedance converter (NIC) circuit is introduced which compensates for the stray capacitance at the output node due to the cable connecting the current driver to the electrodes. Finally, a new current driver in 0.6-μm CMOS technology is presented using feedback to provide high output impedance from DC to over1 MHz. Simulation results including Monte-Carlo analysis are presented to show the operation of the circuit. <s> BIB005 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> Current drivers are fundamental circuits in bioimpedance measurements including electrical impedance tomography (EIT). In the case of EIT, the current driver is required to have a large output impedance to guarantee high current accuracy over a wide range of load impedance values. This paper presents an integrated current driver which meets these requirements and is capable of delivering large sinusoidal currents to the load. The current driver employs a differential architecture and negative feedback, the latter allowing the output current to be accurately set by the ratio of the input voltage to a resistor value. The circuit was fabricated in a 0.6-μ m high-voltage CMOS process technology and its core occupies a silicon area of 0.64 mm2. It operates from a ± 9 V power supply and can deliver output currents up to 5 mA p-p. The accuracy of the maximum output current is within 0.41% up to 500 kHz, reducing to 0.47% at 1 MHz with a total harmonic distortion of 0.69%. The output impedance is 665 kΩ at 100 kHz and 372 k Ω at 500 kHz. <s> BIB006 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> This paper describes an alternative sinusoidal current driver suitable for bioimpedance applications where high frequency operation is required. The circuit is based on a transconductor and provides current outputs with low phase error for frequencies around its pole frequency. This extends the upper frequency operational limit of the current driver. Multifrequency currents can be generated where each individual frequency is phase corrected. Analysis of the circuit is presented together with simulation and experimental results which demonstrate the proof of concept for both single and dual frequency current drivers. Measurements on a discrete test version of the circuit demonstrate a phase reduction from 25 $^{\circ}$ to 4 $^{\circ}$ at 3 MHz for 2 mAp-p output current. The output impedance of the current driver is essentially constant at about 1.1 M $\Omega$ over a frequency range of 100 kHz to 5 MHz due to the introduction of the phase compensation. The compensation provides a bandwidth increase of a factor of about six for a residual phase delay of 4 $^{\circ}$ . <s> BIB007 </s> Analog Integrated Current Drivers for Bioimpedance Applications: A Review <s> Closed Loop Linear Feedback Integrated Current Drivers <s> Bioimpedance measurements are performed in a variety of medical applications including cancer detection in tissue. Such applications require wideband (typically 1 MHz) accurate ac current drivers with high output impedance and low distortion. This paper presents an integrated current driver that fulfills these requirements. The circuit uses negative feedback to accurately set the output current amplitude into the load. It was fabricated in a 0.35-μm complementary metal–oxide–semiconductor (CMOS) process technology, occupies a core area of 0.4 mm2 , and operates from ±2.5-V power supplies. For a maximum output current of 1 mAp-p, the measured total harmonic distortion is below 0.1%, and the variability of the output current with respect to the load is below 0.5% up to 800 kHz increasing to 0.86% at 1 MHz. The current driver was tested for the detection of cancer sites from postoperative human colon specimens. The circuit is intended for use in active electrode applications. <s> BIB008
The linear feedback current driver in BIB004 is based on a linear feedback system. The schematic is shown in Figure 7 . 8 to 1 MHz provided 0.8% current variations where the maximum operating current was 500 μAp-p BIB002 . The output impedance at 10 kHz was >1 MΩ, however, it was reduced to >160 kHz at 1 MΩ frequency. The THD of the maximum output current was 42 dB (0.79%), measured at 600 kHz BIB002 . The linear feedback current driver in BIB004 is based on a linear feedback system. The schematic is shown in Figure 7 . Figure 7 . Basic linear feedback current driver BIB004 . The current driver in this architecture employs a high transconductance OTA (G) in a negative feedback configuration to sense and regulate the current through the load. The generated output current flows through the load resistor (RL) that is in series with the sense resistor (Rs). The resulting voltage controls the difference between Vin and Vi, thus monitoring and modifying changes in the output current. The output current of the driver is: where (Gm) is the transconductance, (ro) is the output resistance of the transconductor, (RL) is the load resistance, and (Rs) is the sense resistance. Equation  illustrates the dependency of output current Figure 7 . Basic linear feedback current driver BIB004 . The current driver in this architecture employs a high transconductance OTA (G) in a negative feedback configuration to sense and regulate the current through the load. The generated output current flows through the load resistor (R L ) that is in series with the sense resistor (R s ). The resulting voltage controls the difference between V in and V i , thus monitoring and modifying changes in the output current. The output current of the driver is: where (G m ) is the transconductance, (r o ) is the output resistance of the transconductor, (R L ) is the load resistance, and (R s ) is the sense resistance. Equation  illustrates the dependency of output current upon the OTA's internal parameters (G m , r o ), the load resistance (R L ), and the sense resistance (R s ). For r o >> R L + R s and G m R s >>1, the equation then reduces to The overall transconductance of the circuit is only dependent on the passive component R s , which can be accurately trimmed in integrated circuits and is not a function of circuits internal parameters. The output impedance of the current driver is: As observed in Equation (14), the output resistance of the current driver is enhanced through the negative feedback configuration by multiplication of the OTA's open loop gain factor (G m r o ) with the sense resistor. Hence, the OTA's output stage is simply increased without any design techniques to reach the output resistance in the MΩ region for accurate current measurements. The utilization of negative feedback increases the operating bandwidth of the driver. Amplifiers with high open loop gain are not crucial in the design. Nevertheless, this current driver suffers from two major drawbacks. The first problem arises due to voltage imbalance across load. This is the outcome of the voltage drop across the sense resistor as a result of output current passing through it BIB003 . This voltage imbalance results in common mode voltages that can be troublesome in bioimpedance measurements. Common mode voltages, due to imbalances, are at the same frequency as differential voltages, and their suppression is limited due to the degradation of the amplifier's CMRR at higher frequencies BIB003 . The presence of common mode signals can result in differential signal measurement errors, which can lead to false impedance estimations. Finally, common mode voltage errors in bioimpedance measurements can sometimes be higher than differential voltages at the input of the measuring differential amplifiers due to the high input impedance BIB003 . Another disadvantage of this topology is the need to employ floating input voltages. The application of a floating input isolates the load from any direct path to instrument ground. In addition, it serves as a reference to the inverting terminal of the OTA when the voltage across the sense resistor changes BIB003 . The measured voltage developed across the sense resistor should provide a direct indication of the load current. Hence, shunting the load current will affect the accuracy in the regulation of the output current. Although transformer coupling can be utilized to generate balanced floating AC inputs, this is impractical to implement in CMOS. The transconductor was implemented by using two cascaded OTAs to achieve large transconductance, as shown in Figure 8 BIB004 . Transistors M1A, M1B, M2A, and M2B form a source degenerated voltage to current converter. M3A and M3B operate as active loads, where M4A and M4B provide feedback to stabilize the common mode at the output. The overall transconductance of the OTA is given by: where Gm2 (= Gm1), is the transconductance of the second OTA. The simulation result shows an output impedance of 10 MΩ up to 10 kHz, which degraded to 1 MΩ at 1 MHz frequency with the presence of a 1 kΩ load BIB004 . With a supply voltage of ±2.5 V, the drive current capability is up to 500 μAp-p. The THD for the maximum current is -67 dB at 10 kHz and degrades to −52 dB at 1 MHz BIB005 . When frequency effects are considered, the transconductance (G) normally has two poles, one of which is dominant to ensure stability. Although the finite loop gain starts to roll off at cut-off frequency and degrade the output resistance to some extent, the effect of finite bandwidth can be modeled as the output resistance Rout in parallel to a capacitor Cout. Therefore, the open loop frequency response is given by: Where ω = 1/RoutCout. Considering Equation (14), due to low frequency dominant pole ωd in the Transistors M1A, M1B, M2A, and M2B form a source degenerated voltage to current converter. M3A and M3B operate as active loads, where M4A and M4B provide feedback to stabilize the common mode at the output. The overall transconductance of the OTA is given by: where G m2 (= G m1 ), is the transconductance of the second OTA. The simulation result shows an output impedance of 10 MΩ up to 10 kHz, which degraded to 1 MΩ at 1 MHz frequency with the presence of a 1 kΩ load BIB004 . With a supply voltage of ±2.5 V, the drive current capability is up to 500 µA p-p . The THD for the maximum current is -67 dB at 10 kHz and degrades to −52 dB at 1 MHz BIB005 . When frequency effects are considered, the transconductance (G) normally has two poles, one of which is dominant to ensure stability. Although the finite loop gain starts to roll off at cut-off frequency and degrade the output resistance to some extent, the effect of finite bandwidth can be modeled as the output resistance R out in parallel to a capacitor C out . Therefore, the open loop frequency response is given by: BIB007 where ω = 1/R out C out . Considering Equation BIB004 , due to low frequency dominant pole ω d in the two-pole loop gain, the output impedance at high frequencies is not resistive, but approximately capacitive. This capacitor is BIB007 : The value for this capacitor is in order of 2-10 pF, in addition to any added parasitic capacitance. The output impedance is frequency dependent, and at high frequencies it decreases, which results in degraded current accuracy. Moreover, due to the capacitor, the output phase delay increases at higher frequencies and creates phase delay between the input and output of the current driver. There can be a large offset at the output of the high transconductance OTA in Figure 8 , mainly due to the fact that the output DC voltage of each OTA is controlled through the quiescent output resistance of the triode transistors of the corresponding transconductor, provided as common mode feedback for differential output. This quiescent resistance is dependent upon device dimensions. As a result, in the presence of any fabrication inaccuracies or device mismatches, the output DC level cannot be accurately defined, and results in large DC offset. The current driver in BIB006 has been introduced to overcome the two major limitations of floating input and voltage imbalance at the load in BIB004 . The block diagram of the topology with two identical differential feedback current drivers is shown in Figure 9 . There can be a large offset at the output of the high transconductance OTA in Figure 8 , mainly due to the fact that the output DC voltage of each OTA is controlled through the quiescent output resistance of the triode transistors of the corresponding transconductor, provided as common mode feedback for differential output. This quiescent resistance is dependent upon device dimensions. As a result, in the presence of any fabrication inaccuracies or device mismatches, the output DC level cannot be accurately defined, and results in large DC offset. The current driver in BIB006 has been introduced to overcome the two major limitations of floating input and voltage imbalance at the load in BIB004 . The block diagram of the topology with two identical differential feedback current drivers is shown in Figure 9 . The use of two identical differential feedback current drivers operating in a balanced mode minimizes common mode voltage errors across the load (Zload). The difficulty about floating inputs is eliminated by utilization of the two voltage buffers (B1, B2) in the feedback loop, isolating the load from the input signal. As discussed in BIB006 , each driver is composed of a preamplifier stage, followed by a transconductance stage. Current through the load is sensed via sense resistors where the resulting voltage is fed back to the negative terminal of the respective preamplifier, thus establishing a negative feedback loop. The transconductance of the circuit is BIB006 : where Iout is the output current of the driver, Vin is the input voltage, Rs is the sense resistor, A is the gain of the preamplifier stage, Gm is the small signal gain, and ro is the small signal output resistance of the transconductance stage. If >> + and >> 1, then Figure 9 . High power current driver BIB006 . The use of two identical differential feedback current drivers operating in a balanced mode minimizes common mode voltage errors across the load (Z load ). The difficulty about floating inputs is eliminated by utilization of the two voltage buffers (B 1 , B 2 ) in the feedback loop, isolating the load from the input signal. As discussed in BIB006 , each driver is composed of a preamplifier stage, followed by a transconductance stage. Current through the load is sensed via sense resistors where the resulting voltage is fed back to the negative terminal of the respective preamplifier, thus establishing a negative feedback loop. The transconductance of the circuit is BIB006 :  As a result, the transconductance of the current drive is only dependent on the sense resistor, not the circuit's internal parameters. The total transconductance is twice the transconductance of a single drive, but the total output impedance is halved, since the two current drivers are in parallel BIB006 . The output impedance is enhanced through the negative feedback as follows: In the frequency range of 10 kHz-500 kHz the accuracy of the output current is 0.41% for maximum current of 2.5 mA. For 10 kHz-1 MHz the accuracy for the maximum current is 0.47% BIB006 . The measured output impedance is 665 kΩ at 100 kHz and 64 kΩ at 1 MHz. The phase delay increased as the frequency was increased. In BIB006 , the circuit can deliver a maximum output current of 5 mA p-p with an accuracy of 0.41% in the frequency range up to 500 kHz, together with enhanced output voltage compliance of 15 V. A THD of less than 1% is achieved for an output current of 5 mA p-p . At 1 MHz frequency the phase delay was 12 • . A major drawback of this system is its stability. Sharp peaking in the frequency response of the system is the result of insufficient phase margins as the frequency approaches its upper limit. This excess phase delay, which is the result of two closely placed poles of transconductor and buffer, is compensated by a 60 pF on-chip capacitor for each driver. The compensation capacitor at the output of the preamplifier stage moves the gain crossover point more towards the origin. The gain-crossing point reaches unity well before the phase crossing point reaches −180 • and provides enough phase margin for stability. Although this approach retains the gain and output swing, it reduces the bandwidth BIB001 . Considering the frequency effects in the two pole loop gain system (AG m r o R s ) according to BIB006 , due to presence of low frequency dominant pole (ω d ) of the system, the output impedance at high frequencies is not resistive but approximately acts as a capacitor BIB007 . Therefore, as the frequency is increased, the phase delay introduced as the result of the capacitive impedance increases. This phase delay between the input and output directly affects any impedance measurement accuracy. Another limitation of this design is the increased power consumption required in comparison to other CMOS designs, as using two current driver circuits operating in parallel for the purpose of load voltage balancing doubled the total power consumption. Although the model for two parallel current drivers was established for balancing the output voltage, in reality, some DC offset is present at the output. This is mainly due to the fact that the output DC voltage of each driver is controlled through the quiescent output resistance of the triode transistors of the corresponding transconductor provided as common mode feedback for differential output transconductor. This quiescent resistance is dependent upon device dimensions. As a result, in the presence of any fabrication inaccuracies or device mismatches, the output DC level cannot be accurately defined and results in a large DC offset. In BIB008 , a similar architecture has been used in a 0.35 µm standard CMOS technology that operates with a ±2.5 V supply. The schematic of the current driver is shown in Figure 10 .