1. Field of the Invention
The present invention relates to a predistortion compensation apparatus for performing distortion compensation processing in advance to a transmission signal before amplification.
2. Description of the Related Art
In recent years, high-efficient digital transmission has been adopted in the radio communication field. When multilevel phase modulation is adopted in the radio communication, a technique for reducing adjacent channel leak power becomes important, in which nonlinear distortion is restrained by linearizing the amplification characteristic of a power amplifier on the transmission side.
Also, to improve power efficiency even in case an amplifier having a degraded linearity is used, a technique for compensating nonlinear distortion for the degraded linearity is necessary.
FIG. 1 shows an exemplary block diagram of transmission equipment in the conventional radio equipment. A transmission signal generator 1 outputs a digital serial data sequence. Also, a serial-to-parallel (S/P) converter 2 converts the digital data sequence into two series, in-phase component (I-component) signals and quadrature component (Q-component) signals, by alternately distributing the digital data sequence on a bit-by-bit basis.
A digital-to-analog (D/A) converter 3 converts the respective I-signals and Q-signals into analog baseband signals, and inputs the signals into a quadrature modulator 4. This quadrature modulator 4 performs orthogonal transformation and outputs signals by multiplying the input I-signals and Q-signals (transmission baseband signals) by a reference carrier wave 8 and a carrier wave phase-shifted therefrom by 90°, respectively, and adding the multiplied results.
A frequency converter 5 mixes the quadrature modulation signals with local oscillation signals, and converts the mixed signals into radio frequency. A transmission power amplifier 6 performs power amplification of the radio frequency signals output from frequency converter 5, and radiates the signal to the air from an antenna 7.
Here, in the mobile communication using W-CDMA, etc., transmission equipment power is substantially large, becoming as much as 10 mW to several tens of mW, and transmission power amplifier 6 has a nonlinear input/output characteristic having a distortion function f(p), as shown by the dotted line in FIG. 2. This non-linearity causes a non-linear distortion. As shown by the solid line (b) in FIG. 3, the frequency spectrum in the vicinity of a transmission frequency f0 comes to have a raised sidelobe from the characteristic shown by the broken line (a). This leaks to adjacent channels and produces adjacent interference. Namely, due to the nonlinear distortion shown in FIG. 2, leak power of the transmission wave to the adjacent frequency channels becomes large, as shown in FIG. 3.
An ACPR (adjacent channel power ratio) is used to indicate the magnitude of leak power. ACPR is a ratio of leak power to adjacent channels to the power in the channel of interest, in other words, a ratio of the spectrum area in the adjacent channels sandwiched between the lines B and B′ in FIG. 3 to the spectrum area between the lines A and A′. Such leak power affects other channels as noise, and degrades communication quality of the channels concerned. Therefore, a strict regulation has been established to the issue of leak power.
The leak power is substantially small in a linear region of, for example, a power amplifier (refer to a linear region I in FIG. 2), but is large in a nonlinear region II. Accordingly, to obtain a high-output transmission power amplifier, the linear region I has to be widened. However, for this purpose, it becomes necessary to provide an amplifier having a larger capacity than is actually needed, which causes disadvantage in apparatus cost and size. As a measure to solve this problem, a distortion compensation function to compensate for transmission power distortion is added to radio equipment.
FIG. 4 shows the block diagram of transmission equipment having a digital nonlinear distortion compensation function by use of a DSP (digital signal processor). A digital data group (transmission signals) transmitted from transmission signal generator 1 is converted into two series, I-signals and Q-signals, in S/P converter 2, and then the two series of signals are input to a distortion compensator 9.
As shown in the lower part of FIG. 4 in enlargement, distortion compensator 9 includes a distortion compensation coefficient storage 90 for storing a distortion compensation coefficient h(pi) corresponding to the power level pi (i=0-1023) of a transmission signal x(t); a predistortion portion 91 for performing a distortion compensation process (predistortion) onto the transmission signal, using the distortion compensation coefficient h(pi) corresponding to the transmission signal power level; and a distortion compensation coefficient calculator 92 for comparing the transmission signal x(t) with a demodulation signal (a feedback signal) y(t) demodulated in the quadrature detector which will be described later, and calculates and updates the distortion compensation coefficient h(pi) so that the difference between the transmission signal and the demodulation signal becomes zero.
The signal to which distortion process is performed in distortion compensator 9 is input into D/A converter 3. D/A converter 3 converts the input I-signal and Q-signal into analog baseband signals, and inputs the converted signals into quadrature modulator 4. Quadrature modulator 4 performs quadrature modulation by multiplying the input I-signal and Q-signal by a reference carrier wave 8 and a carrier wave being phase-shifted from carrier wave 8 by 90°. Quadrature modulator 4 then adds and outputs the multiplied result.
A frequency converter 5 mixes the quadrature modulation signal with a local oscillation signal, and performs frequency conversion. A transmission power amplifier 6 performs power amplification of the radio frequency signal output from frequency converter 5, and radiates the signal to the air by an antenna 7.
A portion of the transmission signal is input to a frequency converter 11 via a directional coupler 10, and input into a quadrature detector 12 after being converted by the above frequency converter 11. Quadrature detector 12 performs quadrature detection by multiplying the input signal by a reference carrier wave, and by a signal which is phase shifted by 90° from the reference signal, respectively. Thus, the baseband I-signal and Q-signal on the transmission side are reproduced, which are then input into an analog-to-digital (A/D) converter 13.
A/D converter 13 converts the input I-signal and Q-signal into digital signals, and inputs into distortion compensator 9. Through the adaptive signal processing, using an LMS (least-mean-square) algorithm, in distortion compensation coefficient calculator 92 of distortion compensator 9, the pre-compensated transmission signal is compared with the feedback signal being demodulated in quadrature detector 12. Then distortion compensator 9 calculates the distortion compensation coefficient h(p1) so as to make the above difference zero. Then, distortion compensator 9 updates the above-obtained coefficient which has been stored in distortion compensation coefficient storage 90. Through the repetition of calculations above, nonlinear distortion in transmission power amplifier 6 is restrained, and adjacent channel leak power is reduced.
By way of example, in the PCT International Publication WO 2003/103163, such a configuration as shown in FIG. 5, in which distortion compensation is performed using the adaptive LMS algorithm, is described as an embodiment of distortion compensator 9 shown in FIG. 4.
In FIG. 5, a multiplier 15a corresponds to a predistortion section 91 shown in FIG. 4, in which a transmission signal x(t) is multiplied by a distortion compensation coefficient hn-1(p). Also, a distortion device 15b having a distortion function f(p) corresponds to a transmission power amplifier 6 shown in FIG. 4.
Further, as to the portion in FIG. 4 including a frequency converter 11, a quadrature detector 12 and an A/D converter 13, in which the output signal being output from transmission power amplifier 15b is feedbacked, a feedback system 15c is shown in FIG. 5.
Moreover, in FIG. 5, a look-up table (LUT) 15e constitutes a distortion compensation coefficient storage 90 shown in FIG. 4. A distortion compensation coefficient calculation section 16 constitutes a distortion compensation coefficient calculation section 92 shown in FIG. 4, which generates an update value of the distortion compensation coefficient stored in look-up table 15e. 
In the distortion compensation apparatus having the configuration shown in FIG. 5, look-up table 15e has a distortion compensation coefficient for canceling the distortion produced in transmission power amplifier 6, namely, distortion device 15b, in a two-dimensional address location corresponding to each discrete power value of the transmission signal x(t).
When the transmission signal x(t) is input, an address generation circuit 15d calculates the power p (=x2(t)) of the transmission signal x(t), and generates an address of one dimensional direction, for example the X-axis direction, which uniquely corresponds to the above-calculated power p (=x2(t)) of the transmission signal x(t). At the same time, address generation circuit 15d obtains a difference ΔP of the power P1 (=x2(t-1)) of the transmission signal x(t-1) of the previous time point (t-1) having been stored in address generation circuit 15d, and generates an address of the other dimensional direction, for example, the Y-axis direction, which uniquely corresponds to the above difference ΔP.
Thus, from address generation circuit 15d, a store location in look-up table 15e, which is specified by the address P in the X-axis direction and the address ΔP in the Y-axis direction, is read out. The readout address is output as address designation information (AR).
Then, a distortion compensation coefficient hn-1(p) stored in the above readout address is read out from look-up table 15e, so as to be used in the distortion compensation processing performed by multiplier 15a. 
Meanwhile, an update value for updating the distortion compensation coefficient having been stored in look-up table 15e is calculated in a distortion compensation coefficient calculation section 16. Namely, distortion compensation coefficient calculation section 16 is constituted of a conjugate complex calculation section 16 and multipliers 15h-15j. A subtractor 15g outputs a difference e(t) between the transmission signal x(t) and the feedback demodulation signal y(t). Multiplier 15i multiplies the distortion compensation coefficient hn-1(p) by y*(t), so as to obtain an output u*(t) (=n-1(p)y*(t)). Multiplier 15h multiplies the difference e(t) being output from subtractor 15g by u*(t). Multiplier 15j multiplies the output of multiplier 15h by a step size parameter μ.
Next, an adder 15k adds the distortion compensation coefficient hn-1(p) to the output μe(t)u*(t) being output from multiplier 15j, and obtains an update value of look-up table 15e. This update value is to be stored in a write address (AW), consisting of the X-axis direction address and the Y-axis direction address, being specified by address generation circuit 15d as the address corresponding to the transmission signal power p (=x2(t)).
Here, the aforementioned write address (AW) is the same address as the readout address (AR). However, because of a calculation time, etc. needed to obtain the update value, the readout address is used as the write address after the readout address is delayed in a delay section 15m. 
Delay portions 15m, 15n, 15p add to the transmission signal x(t), the delay time D, which is the period from the input of the transmission signal x(t) to the feed back decoded signal y(t) input to the subtractor 15g. 
The delay time D being set by the delay portions 15m, 15n, 15p is determined so as to satisfy D=D0+D1, where D0 is the delay time in transmission power amplifier 15b, and D1 is the delay time in feedback system 15c. 
Using the above configuration, the following calculations are performed.hn(p)=hn-1(p)+μe(t)u*(t)e(t)=x(t)−y(t)y(t)=hn-1(p)×(t)f(p)u*(t)=x(t)f(p)=hn-1(p)y*(t)p=|x(t)|2 Here, x, y, f, h, u, e are complex numbers, and * denotes a conjugate complex number.
Through the above calculation processing, the distortion compensation coefficient h(p) is updated so as to minimize the differential signal e(t) between the transmission signal x(t) and the feedbacked demodulation signal y(t). Finally, the value converges to an optimal distortion compensation coefficient, so that the distortion of the transmission power amplifier is compensated.
Now, in the above calculation, the step size parameter μ determines a degree of effect of an error component e(t) between the transmission signal x(t), i.e. the reference signal, and the feedback demodulation signal y(t), i.e. the feedback signal, on the update value of the distortion compensation coefficient. In the conventional system, the value of the step size parameter μ is set to a fixed value.
In the configuration of the distortion compensation apparatus shown in FIG. 5, the inventor of the present invention has observed the output of the distortion compensation apparatus by inputting the outputs of transmission power amplifier 15b, the distortion device, into a spectrum analyzer with sweep frequencies. FIGS. 6A through 7B are the results obtained at those times. In the examples shown in FIGS. 6A through 7B, the observations have been made with different transmission signal levels in four frequency bands (channels).
FIGS. 6A, 6B represent the output spectrum waveforms of the spectrum analyzer when the transmission signal level is large (43 dB). FIG. 6A shows the spectrum waveform when the step size parameter μ is set to 1/1024, while FIG. 6B shows the spectrum waveform when the step size parameter μ is set to 1/16.
Also, FIGS. 7A, 7B represent the output spectrum waveforms of the spectrum analyzer when the transmission signal level is small (27 dB). FIG. 7A shows the spectrum waveform when the step size parameter μ is set to 1/1024, while FIG. 7B shows the spectrum waveform when the step size parameter μ is set to 1/16.
From these FIGS. 6A through 7B, it has been found out that the relation between the transmission signal level and the step size parameter μ produces an effect on the distortion compensation coefficient. In FIGS. 6A, 6B, if the step size parameter μ is set larger when the transmission signal level is large (refer to FIG. 6B), external disturbance (phase rotation, quantization error in an A/D converter, etc.) affects greater, resulting in a larger number of rise pulses being produced. By this, the compensation coefficient tends to diverge at the time of calculating the error.
On the contrary, if the step size parameter μ is set smaller when the transmission signal level is small, a minute error having been detected is canceled, which produces a problem of preventing proper update of the compensation coefficient (refer to FIG. 7A).