1. Field of the Invention
The present invention relates generally to DC to DC converters and specifically to a circuit to recover energies in snubbing capacitors by using auxiliary switch(es) in a single-ended forward converter. More particularly, it relates to such a circuit in single-ended interleaved or non-interleaved forward converters. Most especially, it relates to such a circuit which clamps the peak primary switch voltages to below a predetermined voltage, independent of output power, and properly resets the power transformers, introducing virtually no losses.
2. Description of the Prior Art
The demand for distributed power systems in computer and communications industries has mandated the power conversion from a low dc bus voltage, typically 48 volts, to voltage levels suitable for integrated circuits, such as 5.1 volts and 3.3 volts. In the design of such DC-DC converters, one of the safety requirements for meeting CLASS 3 per IEC Publication 950 is that the peak primary voltage cannot exceed 140 volts between any two points under any operating condition.
A single-ended forward-type pulse width modulated (PWM) converter is a prime candidate for this application. At higher power levels, interleaved type forward converters are used. The converter usually operates with an input voltage range of 40-60 volts for accommodating a possible battery back up. To prevent the peak primary voltage from exceeding 140 volts, a large snubbing capacitance has to be used across the main switch. The energy in the snubbing capacitance is lost when the main switch turns on, and thus the converter's efficiency suffers. As the output power increases, the situation becomes more severe because the energy in the leakage inductance increases and as a result the snubbing capacitance has to be larger, resulting in greater loss of efficiency as the larger snubbing capacitance is discharged through the primary switch.
A distributed power system usually consists of a front-end power stage, which converts typical ac outlet voltage to a safe low dc voltage and performs power factor correction to the ac line. The low dc voltage is then distributed to the loads, where small local converters are used to convert the low dc voltage to proper levels, for example for use with integrated circuits. Such an architecture reduces the hazards of high voltage distribution, provides built-in redundancy over systems in which voltages are centrally generated and distributed, and has better dynamic responses to the loads. The most popular dc voltage selected for low dc voltage distribution systems is 48 volts. Many converter modules converting 48 volts to proper voltage levels for integrated circuits have recently been developed and are commercially available. Various converter topologies have been adopted in the converter modules. The single-ended forward topology with resonant reset, as shown in FIG. 1, is among the most popular. For higher power levels, interleaved single-ended forward converters, as shown in FIG. 2, are used. FIGS. 1A and 2A illustrate some of the waveforms present in the circuits of FIGS. 1 and 2, respectively.
When the main switch of a single-ended forward converter turns off, energies stored in the magnetizing and leakage inductances tend to resonate between the inductances and the output capacitance of the main switch, such as a MOSFET, and generate voltage spikes and high-frequency ringing. Note that a MOSFET effectively has a capacitor C.sub.M between its drain and source terminals, and a diode D.sub.M from its source to drain terminal as illustrated by dotted lines in conjunction with a MOSFET in the insert in FIG. 1. These elements are not explicitly illustrated in the circuit schematics, but are understood to be present by those skilled in the art.
Traditionally, resistance-capacitance (R-C) or resistance-diode-capacitance (R-D-C) snubber circuits are used to provide a low impedance path to limit the spikes caused by high change of current with respect to time (di/dt), and/or to damp the ringing. An R-C snubber circuit is a resistor connected to the MOSFET main switch drain in series with a capacitor connected to the source. An R-D-C snubber circuit has the structure of the R-C snubber circuit but with an additional diode in parallel with the resistor and with its anode connected to the drain and the cathode connected to the capacitor. These types of snubber circuits, however, are very lossy and inefficient.
Another traditional approach is to add only external capacitance across the main switch to decrease the characteristic impedance and the resonant frequency. This results in a reduction of the spikes and the ringing by significantly reducing the resonant frequency. However, the introduction of the additional capacitance disadvantageously increases the turn-on energy losses of the main switches (the capacitor energies are dissipated when the main switch turns on), and the converter's efficiency suffers. To meet CLASS 3 per IEC Publication 950 safety requirements, large capacitances are needed to prevent the peak main switch voltage(s) from exceeding 140 V under all operating conditions. The requirement for large capacitances, typically about 20 to 50 times the output capacitance of the MOSFET, serving as the main switch, depending on the energy of the transformer leakage induction, makes this approach unattractive. Although the spikes and ringing can also be reduced by increasing the magnetizing inductance, there is also a practical limitation, as well as cost concerns, to making large magnetizing inductances.
Another approach, the zero-voltage switched-snubber concept, may effectively clamp the voltage across the main switch or the output rectifier diodes in PWM converters to a specified level while recovering the energies in the snubbing capacitor. The zero-voltage switching concept requires that the switching of the main switch occur when the voltage of the snubbing capacitor is zero or a very low voltage. As the voltage across the main switch at the time of its switching on increases, the required snubbing capacitance increases as the square of the voltage. A larger snubbing capacitance means greater energy loss when switching. The zero-voltage switched snubber concept allows the use of lower voltage rated switching devices as the main switch, thus reducing the devices' conduction losses and improving the overall efficiency of the converter. However, although zero-voltage-switching can be achieved using switched snubbers, such implementation requires additional gate-drive logic and extra drivers for the auxiliary switch or switches. This additional gate-drive logic and driver(s) is not commercially available as a single integrated device, and thus must be made specifically for this purpose, resulting generally in greater size and cost. Furthermore, the transformers having specially designed characteristics may be needed to operate in conjunction with these additional components.
The typical circuit operation of a single-ended forward converter as in FIG. 1 is now described. When MOSFET main switch M1 turns on, capacitor CS1 is totally discharged (V.sub.CS1 =0), diode DR1 is conducting, and diode DR2 is off. The majority of the input voltage, E, given by the expression ##EQU1## where: LM1=transformer magnitizing inductance,
LK1=transformer leakage inductance, PA1 E=voltage provided by voltage source E, PA1 N=the primary to secondary transformer turns ratio PA1 E=voltage provided by voltage source E, PA1 V.sub.DR2 =voltage across diode DR2 PA1 i.sub.p =reflected load current in the transformer primary winding, PA1 I.sub.0 =load current in RL, and PA1 N=the primary to secondary transformer turns ratio.
is transformed to the transformer secondary, and the voltage across diode DR2 is given by, EQU V.sub.DR2 .perspectiveto.E/N (2)
where:
and the load current is reflected back to the transformer primary. Therefore, the reflected load current in the primary, i.sub.p, is related to the load current, I.sub.0, by the expression, EQU i.sub.p .perspectiveto.I.sub.0 /N (3)
where:
When main switch M1 is turned off, the current in leakage inductor LK1, designated i.sub.LK1 (which includes the magnetizing current i.sub.LM1, and the reflected load current, i.sub.p) is diverted to capacitor CS1. The voltage V.sub.CS1 across capacitor CS1 rises, and the voltage transformed to the secondary decreases. When V.sub.CS1 exceeds the magnitude of E, negative current flows in the primary inducing voltage in the secondary such that diode DR2 is turned on and diode DR1 starts to be commutated. During the commutation interval, transformer T1 secondary winding is effectively shorted because both diode DR1 and DR2 are conducting (for a very brief time) and inductor current i.sub.LK1 resonates with capacitor voltage V.sub.CS1.
When diode DR1 turns off, output current freewheels through diode DR2 and the capacitor voltage V.sub.CS1 begins to resonate with i.sub.LM1. (At this instant, the magnitizing current i.sub.LM1 is usually negative. In this event, capacitor voltage V.sub.CS1 decreases.) When V.sub.CS1 decreases below E, positive current flows in the primary, inducing a voltage in the secondary and causing diode DR1 to conduct and commutate diode DR2. However, when diode DR1 starts conducting, reflected current i.sub.P forces V.sub.CS1 to increase and try to commutate diode DR1. If diode DR1 is commutated, the negative magnitizing current i.sub.LM1 forces V.sub.CS1 to decrease again. This mechanism forces V.sub.CS1 to resonate with i.sub.LK1 around the input voltage level until MOSFET main switch M1 is turned on (triggered) by signal V.sub.G1 applied to its gate. At that instant, V.sub.CS1 decreases to zero and the energy in capacitor CS1 is dissipated through MOSFET main switch M1. Diode DR1 is forced to conduct and diode DR2 is commutated. Load current I.sub.0 is again reflected to the primary side of transformer T1 until MOSFET main switch M1 is turned off by trigger signal V.sub.G1, which starts another cycle of the operation.