The example embodiments described below relate generally to sensing the current provided by a switching voltage regulator.
Server farms using high-end processors can place large demands on power and cooling resources. Server system control facilities monitor power consumption and in a dynamic mode processors are switched between different performance states to conserve power. Processor power consumption is determined by measuring the processor current.
Voltage regulator (VR) designs incorporate s simple current sensing technique that utilizes the VR inductor series parasitic resistance (RL). The current through the inductor of the VR is practically equal to the processor current when processor current is constant and this technique provides acceptable accuracy during static mode. However, in dynamic mode monitoring inductor current instead of actual processor current does not provide acceptable results because, while the processor current value changes rapidly and spans a wide range, the current through the VR inductor changes slowly due to LC-filtering effects on the inductor current signal. As a result, in dynamic mode the technique of representing actual processor current by VR inductor current produces errors in both timing and current magnitude monitoring.
A DC-to-DC voltage regulator typically is used to convert a DC input voltage to either a higher or a lower DC output voltage. One type of voltage regulator is a switching regulator that is often chosen due to its small size and efficiency. The switching regulator typically includes one or more switches that are rapidly opened and closed to transfer energy between an inductor (a stand-alone inductor or a transformer, as examples) and an input voltage source in a manner that regulates the output voltage.
As an example, referring to FIG. 1, one type of switching regulator is a Buck switching regulator 10 that receives an input DC voltage (called VIN) and converts the VIN voltage to a lower regulated output voltage (called VOUT) that appears at an output terminal 11. To accomplish this, the regulator 10 may include a switch 20 (a metal-oxide-semiconductor field-effect-transistor (MOSFET), for example) that is operated (via a switch control signal called VSW) in a manner to regulate the VOUT voltage, as described below.
Referring also FIGS. 2 and 3, in particular, the switch 20 opens and closes to control energization/de-energization cycles 19 (each having a duration called TS) of an inductor 14. In each cycle 19, the regulator 10 asserts, (drives high, for example) the VSW signal during an on interval (called TON) to close the switch 20 and transfer energy from an input voltage source 9 to the inductor 14. During the TON interval, a current (called IL) of the inductor 14 has a positive slope. During an off interval (called TOFF) of the cycle 19, the regulator 10 deasserts (drives low, for example) the VSW voltage to open the switch 20 and isolate the input voltage source 9 from the inductor 14. At this point, the level of the IL current is not abruptly halted, but rather, a diode 18 begins conducting to transfer energy from the inductor 14 to a bulk capacitor 16 and a load (not shown) that are coupled to the output terminal 11. During the TOFF interval, the IL current has a negative slope, and the regulator 10 may close a switch 21 to shunt the diode 18 to reduce the amount of power that is otherwise dissipated by the diode 18. The bulk capacitor 16 serves as a stored energy source that is depleted by the load, and additional energy is transferred from the inductor 14 to the bulk capacitor 16 during each TON interval.
For the Buck switching regulator, the ratio of the TON interval to the TS interval (i.e., the summation of the TON and TOFF intervals) is called a duty cycle of the regulator and generally governs the ratio of the VOUT voltage to the VIN voltage. Thus, to increase the VOUT voltage, the duty cycle of the regulator may be increased, and to decrease the VOUT voltage, the duty cycle may be decreased.
As an example, the regulator 10 may include a controller 15 (see FIG. 1) that regulates the VOUT voltage by using a current mode control technique. In this manner, the controller 15 may include an error amplifier 23 that amplifies the difference between a reference voltage (called VREF) and a voltage (called VP (see FIG. 1)) that is proportional to the VOUT voltage to produce an error voltage (called VCNTRL) that is used to control the levels of the VOUT voltage and the IL inductor current.
The controller 15 uses the VCNTRL voltage and a voltage (called VCS) that indicates the IL inductor current to produce the VSW switch control signal to control the switch 20. More specifically, referring also to FIG. 5, the controller 15 may include a comparator 26 that compares the VCNTRL and VCS voltages. The VCS voltage is provided by a differential amplifier 24 that senses the voltage difference (called VR) across a current sensing resistor 29 that is coupled in series with the inductor 14.
The output terminal of the comparator 26 may be coupled to a switch circuit 27 that generates the VSW switch control signal. As an example of one type of current mode control, the switch circuit 27 may keep the TOFF time interval constant and use the positive incline of the VCS voltage to control the duration of the TON time interval. Thus, the TON time interval ends when the VCS voltage reaches the VCNTRL voltage and begins at the expiration of the constant TOFF interval.
Due to the above-described arrangement, when the VOUT voltage increases, the VCNTRL voltage decreases and causes the duty cycle of the regulator 10 to decrease to counteract the increase in VOUT. Conversely, when the VOUT voltage decreases, the VCNTRL voltage increases and causes the duty cycle to increase to counteract the decrease in VOUT. When the average value, or DC component, of the IL current increases, the DC component of the VCS voltage increases and causes the duty cycle to decrease to counteract the increase in the IL current. Conversely, when DC component of the IL current decreases, the DC component of the VCS voltage decreases and causes the duty cycle to increase to counteract the decrease in the IL current. The switching frequency (i.e., 1/TS) typically controls the magnitude of an AC ripple component (called VRIPPLE (see FIG. 4)) of the VOUT voltage, as a higher switching frequency typically reduces the magnitude of the VRIPPLE voltage.
Accurate monitoring of High-end processors current provides precise information about power consumption, which allows for optimal server system control and results in the best data center performance. The faster and the more accurate the current-sensing circuit is, the better the processor and server platform performance.
To monitor CPU current (power), present Voltage Regulator (VR) designs incorporate a simple current-sensing technique that utilizes the VR inductor series parasitic resistance, R L. When processor current is constant, the current through the inductor Is practically equal to the CPU current and this technique provides acceptable accuracy. However, in dynamic mode, sensing the inductor current instead of the actual CPU current has one major disadvantage: while the CPU current value changes rapidly and spans a wide range, current through the inductor changes slowly due to LC-filtering effect on the current signal. Consequently, in dynamic mode this technique produces errors in both timing and current magnitude monitoring.