One common type of electrical power converter that produces a regulated output voltage is a switch mode power supply or a switched supply. Conventional switch mode power supplies commonly include a compact power transformer and one or more power switches for alternately coupling a DC voltage across a primary winding of the power transformer, thereby generating a series of voltage pulses across one or more secondary windings of the power transformer. These pulses are then rectified and filtered to provide one or more output DC voltages. The output voltage or voltages of the power converter are commonly regulated by controlling the relative amount of time that the power switch is on (i.e., the duty cycle).
There are several physical effects that tend to limit the maximum power and power conversion efficiency that can be achieved from a switch mode power supply in which the primary current to a transformer is switched. One limitation is that switch-mode power supplies include switching elements and rectifier elements whose resistance may limit their performance at high current levels. As is well known, resistive power losses increase as a function of I.sup.2 R, where I is the current flowing through the device and R is the electrical resistance of the device. In high current switch mode power supplies, the resistance of individual rectifiers and switches may lead to significant resistive losses. Moreover, the series resistance of a forward biased rectifier and the on-resistance of a transistor switch, such as a field effect transistor (FET), includes the resistance of bond wires and component leads, which can be significant.
The efficiency of a switch-mode power supply may also be limited by transformer winding losses. Compact switch mode power supplies typically operate at a high switching frequency. A high switching frequency permits the transformer size to be reduced while maintaining a desired induced voltage across the secondary winding. However, the efficiency of transformers degrades at high frequency because of increased resistive losses in the primary and secondary windings. Classical electromagnetic theory teaches that at high frequency the current distribution in a wire decreases exponentially with a characteristic length, or skin depth, from the surface. The skin depth varies inversely as the square root of the frequency and the conductivity of a metal. For example, at a frequency of 1 MHZ, the skin depth decreases to 66 .mu.m, such that only a small annulus of a wire conducts. The effective cross-sectional area for current flow thus decreases dramatically at high frequency, leading to a corresponding increase in resistance of the primary and secondary windings.
Another limitation to high frequency operation of a low-profile transformer is leakage inductance. The leakage inductance occurs because not all of the magnetic flux generated by the primary winding is coupled by the core to the secondary winding. Some of the magnetic flux generated by the primary winding does not intersect the secondary winding but instead passes through the air space around the sides of the primary and secondary windings. In the equivalent circuit model of a transformer this leakage flux is modeled as a corresponding parasitic leakage inductance that must also be driven by the primary current but which does not couple power to the secondary winding. The transformer leakage inductance thus has the effect of impeding the flow of power from the primary winding to the secondary winding. If the leakage inductance is large, then a large primary winding current is required to provide a given load current. Also, for some switch-mode power supply topologies, a large leakage inductance may cause overvoltages in power switches at switch turn-on and turn-off. For example, it is well known that a transformer's leakage inductance may cause a voltage spike during the turnoff of a MOSFET switch used as the power switch in a forward converter power supply.
Still another effect that limits the potential efficiency of a switch mode power supply is switching losses. Non-ideal switches are commonly modeled as ideal switches with an additional impedance element that experiences a large switching current and voltage for a brief period of time at each switching event. As is well known in the art, switching losses during each switching event depend upon the power dissipation in the turn-on and turn-off phase of each switching event. Mathematically, the power dissipation per switching event can be calculated as the time interval of the product of the current and the voltage across the switch during one switching event. In an ideal switch, the switch turns on and off instantly such that the time integrated product of the current and voltage through the device is zero. However, in real FETs, there is a finite period of time when the product of the current and the voltage is non-negligible.
The time-averaged switching power loss is the product of the switching loss per switching event (i.e., the power loss per turn-on and turn-off event) multiplied by the switching frequency. The switching loss for a compact switch mode power supply operating at a high frequency (e.g., 100 kHz to 2 MHZ) tends to be large. Moreover, the switching losses are exacerbated at high current levels. For many FET switches used in switching power supplies, the switching loss per switching cycle depends upon the source-drain inductance of FET switches, according to the function 1/2 L.sub.SD I.sub.FET.sup.2, where L.sub.SD is the source-drain terminal inductance of the FET, and I.sub.FET is the FET switching current.
One approach to improve the maximum current capability in a switch-mode power supply is to combine the outputs of two or more individual power supply circuits. FIG. 1 is circuit schematic of a prior art forward converter power supply. The general principles of forward converter supplies similar to that shown in FIG. 1 are well known. Input terminals 2, 4 are connected to an unregulated DC voltage source, V.sub.d. A switch 6 controls the current to the primary winding 8 of transformer 10. Switch 6 may comprise any controllable switch, but typically comprises a field effect transistor (FET). The primary winding 8 induces a voltage in secondary winding 12 of transformer 10. A rectifying diode 14 is connected to one of the output terminals 13 of secondary winding 12. A filter 16 is coupled between the output of diode 14 and the ground terminal 11 of the secondary winding 12. The filter typically includes an inductor 20, freewheeling diode 18, and filter capacitor 22 to filter the current to a load 24. The principles of operation of forward converter circuits is well known. Assuming that transformer 10 is ideal, when switch 6 is on, diode 14 becomes forward biased and freewheeling diode 18 becomes reverse biased. Ignoring diode voltage drops, the voltage, V.sub.L, across the inductor 20 is the induced voltage across the secondary winding 12 minus the load voltage. This can be expressed by the equation: V.sub.L =(N.sub.2 /N.sub.1)V.sub.d -V.sub.o, where N.sub.2 /N.sub.1 is the turns ratio of the transformer, V.sub.d is the input voltage on the primary winding 8, and V.sub.o is the load voltage. As is well known, the voltage across an inductor depends upon the time rate of change of current through the inductor, or V.sub.L =LdI.sub.L /dt, where L is the inductance of the inductor and I.sub.L is the current through inductor 20. Consequently, when the switch 6 is turned on the current tends to ramp up linearly in inductor 20. When the switch 6 is off, there is no induced voltage across the secondary winding 12. Diode 14 is non-conducting. Current from inductor 20 flows through freewheeling diode 18 which becomes conducting as a result of a shift in sign of the voltage across inductor 20. When the switch 6 is off (and ignoring voltage drops across freewheeling diode 18) the voltage across the inductor 20 is equal in magnitude and opposite in sign to the load voltage, or V.sub.L V.sub.o. The current across the inductor gradually ramps down when switch 6 is off. The ratio of the load voltage to the input voltage is V.sub.o /V.sub.d =(N.sub.2 /N.sub.1) D, where D is the duty ratio of switch 6.
As illustrated in FIG. 2, the power output of a switch mode power supply, such as a forward converter, can be improved by combining the current outputs of parallel power supply circuits. A first current source 30 comprises a first transformer 32. Current through the primary winding 34 of first transformer 32 is controlled by a first switch 36. The output of secondary winding 38 of first transformer 32 is coupled to rectifying diode 40. A second current source 44 comprises a second transformer 46. Current through primary winding 48 of second transformer 46 is controlled by a second switch 50. The output of secondary winding 52 of second transformer 46 is coupled to a rectifying diode 54. The currents i.sub.1 and i.sub.2 from first current source 30 and second current source 44 are combined in inductive filter element 58. An additional filter capacitor 60 is coupled across the output of inductor 58 to filter the voltage to a load 62. Each current source may have its own freewheeling diode 42, 56 or a common freewheeling diode may be used. The topology of the forward converter of FIG. 2 comprises essentially two individual forward converters similar to that shown in FIG. 1 driving a common inductor and filter capacitor. Typically, the two switches 36, 50 are sequenced to turn-on a halftime period apart from one another. This results in twice an many current pulses entering inductor 58 compared with a single current source. Operating the two circuits in parallel increases the maximum load current compared to the individual forward converter of FIG. 1. However, the parallel circuit power converter of FIG. 2 has several drawbacks. One significant drawback is that two separate transformers 32, 46 are required, which results in a comparatively expensive, heavy, and bulky power supply design. Another drawback is that additional circuit means are required to turn the two switches 36, 50 on a half-time period apart from one another.
Another approach to improve the performance of switch-mode power supplies is the use of parallel rectifiers and/or switching devices. A common practice in power electronics is to use a parallel arrangement of standard size power switching devices, such as bipolar transistors, field effect transistors (FETs), insulated gate field-effect transistors (IGFETs), and insulated gate bipolar transistors (IGBTs) instead of a single large power device, in order to handle the high currents and power dissipations that may otherwise exceed the safety ratings of a single power switching device. Also, it is a common practice in power electronics to use a parallel arrangement of diode rectifiers to handle high currents that would otherwise exceed the safety ratings of a single diode rectifier.
FIG. 3 shows a portion of a forward converter similar to that of FIG. 1 but with parallel switching devices 70, 72 and parallel diodes 14, 15. Parallel switches 70, 72 may reduce the DC switching resistance compared to a single switch 6. However, a significant drawback of the parallel circuit arrangement of FIG. 3 is that use of parallel switching devices 70, 72 may not substantially reduce the switching losses compared to an individual switching device 6. There is no guarantee of equal current sharing between the parallel power devices 70, 72. Typically, FET transistors are not perfectly matched in their DC current-voltage characteristics. Moreover, there are inductances and capacitances associated with a packaged transistor or diode. Consequently, when two switching devices 70, 72 are connected in parallel, one device typically draws a larger operating current than the other one. Also, one device typically switches on/off before the other one. When two or more parallel power devices are switched, the power device that turns on fastest will absorb all of the turn-on losses. Similarly, the switching device 70, 72 that turns off the slowest will absorb all of the turn-off losses. Consequently, the use of parallel switches 70, 72 may not result in a significant decrease in switching losses compared to a single switch 6.
The use of parallel diodes 14, 15 may also not provide the desired benefit. Equal current sharing of parallel diode rectifiers typically does not occur. Manufacturing variances, coupled with heating effects, tends to make one of the diodes 14, 15 have a significantly different turn-on voltage and series resistance than the other diode. One diode usually draws substantially more current than the other diode. However, each diode 14, 15 has a maximum current or power dissipation rating. The current entering inductor 20 must be limited so that the diode 14, 15 drawing the largest current does not exceed a safe rating. Consequently, the maximum current through parallel diodes 14, 15 must be limited to a value substantially less than that which would be possible if diodes 14, 15 equally shared the current.
Generally, previously known techniques to combine a plurality of conventional transformers, rectifiers, and power switches in parallel do not result in a switch mode power supply that is compact, inexpensive, efficient, has low switching losses and is capable of supplying a substantially increased load current.
What is desired is a new power supply design approach to improve the performance of high-frequency switch-mode power supplies.